TWI556545B - Charge control circuit, flyback power conversion system and charging control method - Google Patents

充電控制電路、反激式電源變換系統及充電控制方法 Charging control circuit, flyback power conversion system and charging control method

本發明涉及一種充電控制電路，更具體地講，本發明涉及一種充電控制電路及使用該充電控制電路的反激式電源變換系統及充電控制方法。 The present invention relates to a charging control circuit, and more particularly to a charging control circuit and a flyback power conversion system and a charging control method using the same.

一般來說，AC/DC電源系統通過變壓器來隔離原邊輸入和副邊輸出，為了把輸出電壓的資訊回饋回控制回路，一般是通過光耦把副邊資訊傳給原邊的控制晶片；但是實際上原邊和副邊輸出電壓有很強的關聯，因此可以通過合適的方式，直接通過控制原邊的資訊來控制輸出電壓。在這種情況下，就可以省掉很多元器件，比如TL431、光耦等，這樣整個電源系統成本能夠大大的降低。 Generally, the AC/DC power system isolates the primary input and the secondary output through a transformer. In order to feed back the output voltage information back to the control loop, the secondary side information is generally transmitted to the primary control chip through the optical coupler; In fact, there is a strong correlation between the primary and secondary output voltages, so the output voltage can be controlled directly by controlling the information of the primary side in a suitable manner. In this case, you can save a lot of components, such as TL431, optocoupler, etc., so that the entire power system cost can be greatly reduced.

第1圖是根據現有技術的反激式電源變換系統的簡要示圖。 Fig. 1 is a schematic diagram of a flyback power conversion system according to the prior art.

第1圖所示的反激式電源變換系統包括整流電路110、充電電路120和充電控制電路130。其中，整流電路110對從交流電源輸入的電壓進行橋式整流並將橋式整流後所得的電壓信號Vin輸入到充電電路120。充電電路120包括原邊繞組Np、副邊繞組Nsec以及輔助繞組Naux以及與其各自相連的二極體、電阻以及電容等輔助電路。原邊繞組Np連接到開關Q。開關Q為雙極性接面電晶體(Bipolar Junction Transistor,BJT)，也可以為金屬氧化物半導體場效應管(Metal-Oxide-Semiconductor Field-Effect Transistor,MOSFET)以及絕緣柵雙極性接面電晶體(Insulated Gate Bipolar Transistor,IGBT)。下面以雙極性接面電晶體BJT為例來描述開關Q與其他部件的連接 關係。這裡，原邊繞組Np連接到開關Q的集電極，開關Q的基極連接到充電控制電路130，而該開關Q的發射極連接有電流檢測電阻Rs，電流檢測電阻Rs的另一端接地。副邊繞組Nsec上連接有二極體D1和電容C1，在電容C1兩端作為輸出端為電池充電。在此，Req為輸出線等效電阻，而Vo表示實際為電池充電的充電電壓Vo。電阻R1和電阻R2串聯之後連接到輔助繞組Naux，電阻R1一端連接到輔助繞組Naux和二極體D3的陽極，另一端與電阻R2相連，電阻R2的另一端接地；二極體D3的陰極和與電容C2連接。這裡，電阻R2上的電壓信號作為回饋電壓被輸入到充電控制電路130，例如輸入到充電控制電路130的FB端子，電流檢測電阻Rs上的電壓信號被輸入到充電控制電路130的CS端子。充電控制電路130根據輸入的電阻R2以及電流檢測電阻Rs上的電壓信號在DRV端子處輸出開關控制信號從而對開關Q進行控制。在充電控制電路130中還可包括退磁檢測模組和採樣控制模組。退磁檢測模組根據FB處輸入的回饋電壓來檢測退磁區間並由此輸出退磁信號。當副邊繞組處於退磁區間時，退磁信號為高電平，否則為低電平。採樣控制模組根據退磁檢測模組輸出的退磁信號來控制採樣控制模組。 The flyback power conversion system shown in FIG. 1 includes a rectifier circuit 110, a charging circuit 120, and a charging control circuit 130. The rectifier circuit 110 performs bridge rectification on the voltage input from the AC power source and inputs the voltage signal Vin obtained by the bridge rectification to the charging circuit 120. The charging circuit 120 includes a primary winding Np, a secondary winding Nsec, and an auxiliary winding Naux, and an auxiliary circuit such as a diode, a resistor, and a capacitor connected thereto. The primary winding Np is connected to the switch Q. The switch Q is a Bipolar Junction Transistor (BJT), and may also be a Metal-Oxide-Semiconductor Field-Effect Transistor (MOSFET) and an insulated gate bipolar junction transistor ( Insulated Gate Bipolar Transistor, IGBT). The bipolar junction transistor BJT is taken as an example to describe the connection of the switch Q with other components. relationship. Here, the primary winding Np is connected to the collector of the switch Q, the base of the switch Q is connected to the charge control circuit 130, and the emitter of the switch Q is connected to the current detecting resistor Rs, and the other end of the current detecting resistor Rs is grounded. A diode D1 and a capacitor C1 are connected to the secondary winding Nsec, and the battery is charged as an output terminal at both ends of the capacitor C1. Here, Req is the output line equivalent resistance, and Vo represents the charging voltage Vo actually charging the battery. The resistor R1 and the resistor R2 are connected in series and then connected to the auxiliary winding Naux. One end of the resistor R1 is connected to the anode of the auxiliary winding Naux and the diode D3, the other end is connected to the resistor R2, the other end of the resistor R2 is grounded, and the cathode of the diode D3 is Connected to capacitor C2. Here, the voltage signal on the resistor R2 is input as a feedback voltage to the charge control circuit 130, for example, to the FB terminal of the charge control circuit 130, and the voltage signal on the current sense resistor Rs is input to the CS terminal of the charge control circuit 130. The charge control circuit 130 outputs a switch control signal at the DRV terminal to control the switch Q based on the input resistor R2 and the voltage signal on the current sense resistor Rs. A demagnetization detection module and a sampling control module may also be included in the charge control circuit 130. The demagnetization detection module detects the demagnetization interval based on the feedback voltage input at the FB and thereby outputs a demagnetization signal. When the secondary winding is in the demagnetization interval, the demagnetization signal is high, otherwise it is low. The sampling control module controls the sampling control module according to the demagnetization signal output by the demagnetization detection module.

第2圖示出了第1圖所示的反激式電源變換系統中回饋電壓、退磁信號、採樣控制信號、輸出電流、原邊電流、以及電流檢測電阻上的電壓的波形變化時序圖。 Fig. 2 is a timing chart showing waveform changes of the feedback voltage, the demagnetization signal, the sampling control signal, the output current, the primary current, and the voltage on the current detecting resistor in the flyback power conversion system shown in Fig. 1.

參照第2圖，開關Q1在t_ON區間導通，而在t_OFF區間關斷，Ts表示一個通斷週期，Tdemag表示在開關Q1關斷時的退磁區間。根據第2圖所示的波形圖可以得知，第1圖所示的反激式電源系統在開關Q1導通時，原邊繞組Np的輸出電流Ipri和電流檢測電阻Rs上的電壓Vcs呈線性逐漸上升至各自最高電流Ipri(0)和最高電壓Vcs_peak，由原邊繞組Np和副邊繞組Nsec構成的變壓器儲藏電能，而電阻R2上的電壓(也第1圖中FB點處的電壓，下面將其稱為回饋電壓V_FB)及副邊輸出電流Isec接近於0，並且退磁信號demag和採樣控制信號Sm-sw輸出低電平。在開關Q1關斷時，原邊的輸出電 流Ipri和電流檢測電阻Rs上的電壓Vcs立即變為0，從而變壓器釋放電能，退磁信號demag輸出高電平以表示退磁過程，同時採樣控制信號Sm-sw輸出高電平以進行採樣，回饋電壓V_FB及副邊輸出電流Isec由其最高值呈線性逐漸下降，這裡，Isec(0)表示副邊輸出電流Isec的最高值。當副邊上的退磁過程幾乎完成，也即副邊繞組Nsec的副邊輸出電流Isec幾乎變為零時，採樣控制信號Sm-sw由高電平變為為低電平，採樣結束。 Referring to Fig. 2, the switch Q1 is turned on in the t _ON interval, and turned off in the t _OFF interval, Ts represents an on-off period, and Tdemag represents the demagnetization interval when the switch Q1 is turned off. According to the waveform diagram shown in FIG. 2, in the flyback power supply system shown in FIG. 1, when the switch Q1 is turned on, the output current Ipri of the primary winding Np and the voltage Vcs of the current detecting resistor Rs are linearly gradually. Rising to the respective highest current Ipri(0) and the highest voltage Vcs_peak, the transformer composed of the primary winding Np and the secondary winding Nsec stores electrical energy, and the voltage across the resistor R2 (also the voltage at the FB point in Fig. 1 below) It is referred to as a feedback voltage V _FB ) and a secondary output current Isec is close to 0, and the demagnetization signal demag and the sampling control signal Sm-sw are outputted at a low level. When the switch Q1 is turned off, the output current Ipri of the primary side and the voltage Vcs of the current detecting resistor Rs immediately become 0, so that the transformer discharges the electric energy, and the demagnetization signal demag outputs a high level to indicate the demagnetization process, while sampling the control signal Sm- Sw outputs a high level for sampling, and the feedback voltage V _FB and the secondary output current Isec gradually decrease linearly from the highest value thereof, where Isec(0) represents the highest value of the secondary output current Isec. When the demagnetization process on the secondary side is almost completed, that is, when the secondary side output current Isec of the secondary winding Nsec becomes almost zero, the sampling control signal Sm-sw changes from the high level to the low level, and the sampling ends.

為了將電池的充電電壓(也即副邊繞組Nsec的輸出電壓)調節到額定電壓範圍內，通常需要提取與輸出電壓和/或輸出負載有關的資訊。在如上所述的單端反激式電源系統工作在電流斷續模式(DCM)的情況下，這些資訊能夠通過輔助繞組Naux很好的被提取出來，在開關Q1開通的時刻，變壓器儲能，當開關Q關斷時候，變壓器上儲存的能量釋放到副邊繞組Nsec的輸出端。此時，輔助繞組Naux的電壓基本上很好地映射了輸出電壓Vo，輔助繞組側FB點處的電壓，也即電阻R2上施加的回饋電壓V_FB和輸出電壓Vo的關係如下所示： In order to adjust the charging voltage of the battery (that is, the output voltage of the secondary winding Nsec) to the rated voltage range, it is usually necessary to extract information related to the output voltage and/or the output load. In the case where the single-ended flyback power supply system as described above operates in the current interrupt mode (DCM), this information can be well extracted through the auxiliary winding Naux, and the transformer stores energy at the moment when the switch Q1 is turned on. When the switch Q is turned off, the energy stored on the transformer is discharged to the output of the secondary winding Nsec. At this time, the voltage of the auxiliary winding Naux is substantially well mapped to the output voltage Vo, and the voltage at the FB point of the auxiliary winding side, that is, the relationship between the feedback voltage V _FB applied to the resistor R2 and the output voltage Vo is as follows:

其中：K=R2/(R1+R2)為回饋係數，N1=Na/Ns為變壓器輔助繞組和副邊匝數比，Na表示輔助繞組Naux的匝數，Ns為副邊繞組Nsec的匝數，V_F是輸出二極體D1上的壓降，Io為輸出電流，即對電池充電的充電電流，Req是輸出線等效電阻，Vo為輸出電壓，也即為電池充電的充電電壓。 Where: K=R2/(R1+R2) is the feedback coefficient, N1=Na/Ns is the ratio of the auxiliary winding of the transformer and the secondary side turns, Na is the number of turns of the auxiliary winding Naux, and Ns is the number of turns of the secondary winding Nsec. V _F is the voltage drop across the output diode D1, Io is the output current, which is the charging current for charging the battery, Req is the equivalent resistance of the output line, and Vo is the output voltage, which is the charging voltage for charging the battery.

第1圖所示的充電控制電路中採用了“採樣保持”的方式：當副邊上的退磁過程幾乎完成，也即副邊繞組Nsec的副邊輸出電流I_sec幾乎變為零時，與輔助繞組Naux的電壓V_aux相應的回饋電壓V_FB例如在第2圖的點A處被採樣，採樣的電壓通常被保持直到下一次退磁開始。同時通過負反饋環路調節採樣電壓到固定的參考電壓Vref，從而獲得和傳統副邊檢測相媲美的恒定電壓的調整率。因此： The charging control circuit shown in Fig. 1 adopts the method of "sampling and holding": when the demagnetization process on the secondary side is almost completed, that is, when the secondary side output current I _sec of the secondary winding Nsec becomes almost zero, The corresponding voltage V _FB of the voltage V _{aux of the} winding Naux is sampled, for example, at point A of Figure 2, and the sampled voltage is typically held until the next demagnetization begins. At the same time, the sampling voltage is adjusted to a fixed reference voltage Vref through a negative feedback loop, thereby obtaining a constant voltage adjustment rate comparable to that of the conventional secondary side detection. therefore:

所以： and so:

由第2圖同時可以知道，輸出電流Io等於副邊輸出電流Isec的平均值，運算式如下： 其中一個通斷週期Ts的副邊輸出電流Isec平均電流Isec_avg為： 其中Isec(0)為Isec的峰值電流；當每一個工作週期副邊輸出電流Isec都相同時，則輸出電流Io等於Isec_avg，即 由變壓器原副邊電流關係可以得到：I sec(0)=N．Ipri(0)因此： It can be seen from Fig. 2 that the output current Io is equal to the average value of the secondary side output current Isec, and the operation formula is as follows: The secondary side output current Isec average current Isec_avg of one of the on-off periods Ts is: Where Isec(0) is the peak current of Isec; when the secondary side output current Isec is the same for each duty cycle, the output current Io is equal to Isec_avg, ie The current relationship between the primary and secondary sides of the transformer can be obtained: I sec(0)= N . Ipri (0) therefore:

其中：N表示原邊繞組Np的匝數和副邊繞組Nsec的匝數的比值，Vcs_peak=Ipri(0)．Rs為CS端子(也即電流檢測電阻Rs)上輸入的電壓峰值，Ipri(0)為原邊峰值電流，Rs為電流檢測電阻。由公式(7)可以知道，可以設定Vcs_peak為定值，Tdemag/Ts為定值來實現恒流輸出。 Where: N represents the ratio of the number of turns of the primary winding Np to the number of turns of the secondary winding Nsec, Vcs _ peak = Ipri (0). Rs is the voltage peak input to the CS terminal (that is, the current detecting resistor Rs), Ipri(0) is the primary peak current, and Rs is the current detecting resistor. It can be known from equation (7) that Vcs_peak can be set to a fixed value, and Tdemag/Ts is a constant value to achieve constant current output.

但是在原邊反激式控制方式的系統中，無論採用單純的恒壓控制還是單純的恒流控制，都存在充電時間長，電池壽命短的問題。因此 需要一種新的控制方式來解決這些問題。 However, in the system of the primary side flyback control method, whether it is simple constant voltage control or simple constant current control, there is a problem that the charging time is long and the battery life is short. therefore A new way of controlling is needed to solve these problems.

兩段式恒流加兩段式恒壓的充電曲線是電池尤其是鎳鉻電池所需要的最優化的充電曲線，根據電池特性所需要的優化充電曲線，提出了本發明所示的充電控制電路及其包含該充電控制電路的反激式電源變換系統及其充電控制方法。本發明在電池充電的過程中採用了兩段式恒流加兩段式恒壓的全新控制方式，從而能夠實現對電池的快速充電並延長電池的壽命。 The two-stage constant current plus two-stage constant voltage charging curve is an optimized charging curve required for a battery, especially a nickel-chromium battery. According to the optimized charging curve required for battery characteristics, the charging control circuit shown in the present invention and the same are provided. A flyback power conversion system of a charging control circuit and a charging control method thereof. The invention adopts a two-stage constant current plus two-stage constant voltage new control mode in the process of charging the battery, thereby realizing rapid charging of the battery and prolonging the life of the battery.

根據本發明的一方面，提供了一種充電控制電路，所述充電控制電路包括：模式選擇電路，接收第一輸入電壓信號，並基於第一輸入電壓信號選擇第一恒流充電控制模式、第一恒壓充電控制模式、第二恒流充電控制模式以及第二恒壓充電控制模式中的一種充電控制模式，並輸出與選擇的充電控制模式有關的第一控制信號以及電壓控制信號；過電流保護電路，接收第二輸入電壓信號和從模式選擇電路輸出的電壓控制信號，並將第二輸入電壓信號與電壓控制信號、第一過電流保護閾值電壓、第二過電流保護閾值電壓進行比較以輸出第二控制信號；邏輯驅動電路，基於從模式選擇電路輸出的第一控制信號以及從過電流保護電路輸出的第二控制信號輸出第三控制信號。 According to an aspect of the present invention, a charge control circuit is provided, the charge control circuit comprising: a mode selection circuit, receiving a first input voltage signal, and selecting a first constant current charge control mode based on the first input voltage signal, first a charging control mode of the constant voltage charging control mode, the second constant current charging control mode, and the second constant voltage charging control mode, and outputting a first control signal and a voltage control signal related to the selected charging control mode; overcurrent protection The circuit receives the second input voltage signal and the voltage control signal output from the mode selection circuit, and compares the second input voltage signal with the voltage control signal, the first overcurrent protection threshold voltage, and the second overcurrent protection threshold voltage to output a second control signal; a logic driving circuit that outputs a third control signal based on the first control signal output from the mode selection circuit and the second control signal output from the overcurrent protection circuit.

根據本發明的另一方面，模式選擇電路包括：退磁檢測器，將第一輸入電壓與第一參考電壓相比較並輸出退磁信號，其中，當退磁過程正在進行時，退磁信號為高電平，當退磁過程結束時，退磁信號為低電平；第一恒壓充電控制電路，基於第一輸入電壓、第二參考電壓和退磁信號輸出第一電平信號和第一電壓信號；第一恒流充電控制電路，基於退磁信號輸出第二電平信號；第二恒流充電控制電路，基於退磁信號輸出第三電平信號，其中，第二電平信號的頻率高於第三電平信號的頻率；第二恒 壓充電控制電路，基於第一輸入電壓和第三參考電壓輸出第四電平信號；控制電壓輸出電路，基於從第一恒壓充電控制電路輸出的第一電壓信號而輸出電壓控制信號；其中，第二電平信號與第四電平信號進行邏輯與操作，並將該邏輯與操作的結果與第一電平信號進行邏輯與操作獲得第一邏輯與操作結果；第三電平信號與第四電平信號進行邏輯與操作，並將該邏輯與操作的結果與第一邏輯與操作結果進行邏輯或操作以輸出第一控制信號。 According to another aspect of the present invention, a mode selection circuit includes: a demagnetization detector that compares a first input voltage with a first reference voltage and outputs a demagnetization signal, wherein the demagnetization signal is at a high level when a demagnetization process is in progress, When the demagnetization process ends, the demagnetization signal is low; the first constant voltage charging control circuit outputs the first level signal and the first voltage signal based on the first input voltage, the second reference voltage, and the demagnetization signal; the first constant current a charging control circuit that outputs a second level signal based on the demagnetization signal; and a second constant current charging control circuit that outputs a third level signal based on the demagnetization signal, wherein the frequency of the second level signal is higher than the frequency of the third level signal Second constant a voltage charging control circuit that outputs a fourth level signal based on the first input voltage and the third reference voltage; and a control voltage output circuit that outputs a voltage control signal based on the first voltage signal outputted from the first constant voltage charging control circuit; The second level signal and the fourth level signal perform a logical AND operation, and logically AND operate the result of the logic and operation with the first level signal to obtain a first logic and operation result; the third level signal and the fourth level The level signal performs a logical AND operation and logically ORs the result of the logic and operation with the first logic and operation result to output a first control signal.

根據本發明的另一方面，第一恒壓充電控制電路包括：採樣控制器，根據退磁檢測器輸出的退磁信號來生成用於控制採樣開關的通斷的採樣控制信號，其中，採樣開關一端接入第一輸入電壓信號，另一端連接至第一電容和第一誤差放大器，第一電容的另一端接地，其中，將第一電容上的電壓作為採樣電壓輸入至第一誤差放大器；第一誤差放大器，將採樣電壓與第二參考電壓之間的差值進行放大以輸出第一電壓信號，並將第一電壓信號輸入至第一比較器和控制電壓輸出電路；斜坡信號發生器，在退磁信號變為高電平的時刻，將斜坡信號重定到第一電壓值，並輸出電壓在第一電壓值和第二電壓值之間逐漸降低的斜坡信號，其中，第一電壓值大於第二電壓值；第一比較器，將從第一誤差放大器輸出的第一電壓信號和斜坡信號發生器輸出的斜坡信號進行比較，並輸出第一電平信號。 According to another aspect of the present invention, a first constant voltage charging control circuit includes: a sampling controller that generates a sampling control signal for controlling on and off of the sampling switch according to a demagnetization signal output from the demagnetization detector, wherein the sampling switch is connected at one end a first input voltage signal, the other end is connected to the first capacitor and the first error amplifier, the other end of the first capacitor is grounded, wherein the voltage on the first capacitor is input as a sampling voltage to the first error amplifier; the first error An amplifier amplifying a difference between the sampling voltage and the second reference voltage to output a first voltage signal, and inputting the first voltage signal to the first comparator and the control voltage output circuit; and the ramp signal generator at the demagnetization signal At a time when the level is high, the ramp signal is reset to the first voltage value, and the ramp signal whose voltage gradually decreases between the first voltage value and the second voltage value is output, wherein the first voltage value is greater than the second voltage value a first comparator that compares the first voltage signal output from the first error amplifier with the ramp signal output from the ramp signal generator And outputting a first level signal.

根據本發明的另一方面，在第一恒流充電控制電路中，第一電流鏡、第一開關、第二開關、第二電流鏡依次串聯連接，退磁信號經過反閘控制第一開關，並直接控制第二開關，第二電容與第二開關和第二電流鏡所形成的串聯電路並聯，第二電容上的電壓信號輸入第二比較器以將該電壓信號與第四參考電壓進行比較並輸出第二電平信號，其中，第二電流鏡的輸出電流是第一電流鏡的輸出電流的第一倍數。 According to another aspect of the present invention, in the first constant current charging control circuit, the first current mirror, the first switch, the second switch, and the second current mirror are sequentially connected in series, and the demagnetization signal passes through the reverse gate to control the first switch, and Directly controlling the second switch, the second capacitor is connected in parallel with the series circuit formed by the second switch and the second current mirror, and the voltage signal on the second capacitor is input to the second comparator to compare the voltage signal with the fourth reference voltage and And outputting a second level signal, wherein an output current of the second current mirror is a first multiple of an output current of the first current mirror.

根據本發明的另一方面，在第二恒流充電控制電路中，第三電流鏡、第三開關、第四開關、第四電流鏡依次串聯連接，退磁信號經過反閘控制第三開關，並直接控制第四開關，第三電容與第四開關和第四電 流鏡所形成的串聯電路並聯，第三電容上的電壓信號輸入第三比較器以將該電壓信號與第四參考電壓進行比較並輸出第三電平信號，其中，第四電流鏡的輸出電流是第三電流鏡的輸出電流的第二倍數，其中，第二倍數大於第一倍數。 According to another aspect of the present invention, in the second constant current charging control circuit, the third current mirror, the third switch, the fourth switch, and the fourth current mirror are sequentially connected in series, and the demagnetization signal passes through the reverse gate to control the third switch, and Directly controlling the fourth switch, the third capacitor and the fourth switch and the fourth battery The series circuit formed by the flow mirror is connected in parallel, and the voltage signal on the third capacitor is input to the third comparator to compare the voltage signal with the fourth reference voltage and output a third level signal, wherein the output current of the fourth current mirror Is the second multiple of the output current of the third current mirror, wherein the second multiple is greater than the first multiple.

根據本發明的另一方面，在第二恒壓控制電路中，第三參考電壓和採樣電壓輸入至第二誤差放大器以將第三參考電壓和採樣電壓之間的差值進行放大並輸出第二電壓信號，第二誤差放大器的輸出端連接有第四電容和壓控振盪器，其中第四電容的另一端接地，壓控振盪器根據第二誤差放大器輸出的電壓信號來輸出第四電平信號。 According to another aspect of the present invention, in the second constant voltage control circuit, the third reference voltage and the sampling voltage are input to the second error amplifier to amplify a difference between the third reference voltage and the sampling voltage and output a second a voltage signal, the output of the second error amplifier is connected to the fourth capacitor and the voltage controlled oscillator, wherein the other end of the fourth capacitor is grounded, and the voltage controlled oscillator outputs the fourth level signal according to the voltage signal output by the second error amplifier. .

根據本發明的另一方面，當採樣電壓小於第三參考電壓時，第四電平信號為高電平；當採樣電壓小於第二參考電壓時，第一電平信號為高電平，第二電平信號的頻率大於第三電平信號的頻率，第一控制信號由第二電平信號決定，模式選擇器選擇第一恒流充電控制模式；當採樣電壓等於第二參考電壓時，第一電平信號為具有第一頻率的電平信號，其中，第一頻率低於第二電平信號的頻率並高於第三電平信號的頻率，第一控制信號由第一電平信號決定，模式選擇器選擇第一恒壓充電控制模式；當採樣電壓大於第二參考電壓小於第三參考電壓時，第一電平信號為低電平，第一控制信號由第三電平信號決定，模式選擇器選擇第二恒流充電控制模式；當採樣電壓等於第三參考電壓時，第一電平信號為低電平，第四電平信號的頻率低於第三電平信號的頻率，第一控制信號由第四電平信號決定，模式選擇器選擇第二恒壓充電控制模式。 According to another aspect of the present invention, when the sampling voltage is less than the third reference voltage, the fourth level signal is at a high level; when the sampling voltage is less than the second reference voltage, the first level signal is at a high level, and the second The frequency of the level signal is greater than the frequency of the third level signal, the first control signal is determined by the second level signal, the mode selector selects the first constant current charging control mode; when the sampling voltage is equal to the second reference voltage, the first The level signal is a level signal having a first frequency, wherein the first frequency is lower than a frequency of the second level signal and higher than a frequency of the third level signal, and the first control signal is determined by the first level signal, The mode selector selects a first constant voltage charging control mode; when the sampling voltage is greater than the second reference voltage is less than the third reference voltage, the first level signal is a low level, and the first control signal is determined by the third level signal, the mode The selector selects a second constant current charging control mode; when the sampling voltage is equal to the third reference voltage, the first level signal is low, and the frequency of the fourth level signal is lower than the frequency of the third level signal, A fourth control signal is determined by the level signal, the mode selector selects the second control mode is the constant voltage charging.

根據本發明的另一方面，第一恒壓充電控制電路輸出的第一電壓信號輸入至控制電壓輸出電路中串聯連接的第一電阻和第二電阻，第二電阻上的電壓作為電壓控制信號經控制電壓輸出電路中的低通濾波器濾波後輸出至過電流保護電路。 According to another aspect of the present invention, the first voltage signal outputted by the first constant voltage charging control circuit is input to the first resistor and the second resistor connected in series in the control voltage output circuit, and the voltage on the second resistor is used as a voltage control signal. The low-pass filter in the control voltage output circuit is filtered and output to the overcurrent protection circuit.

根據本發明的另一方面，過電流保護電路包括：第四比較 器，將第二輸入電壓信號和電壓控制信號相比較，從而在第二輸入電壓信號的電壓等於電壓控制信號的電壓時輸出高電平；第五比較器，將第二輸入電壓信號和第一過電流保護閾值電壓相比較，從而在第二輸入電壓信號的電壓等於第一過電流保護閾值電壓時輸出高電平；第六比較器，將第二輸入電壓信號和第二過電流保護閾值電壓相比較，從而在第二輸入電壓信號的電壓等於第二過電流保護閾值電壓時輸出高電平；其中，第四比較器的輸出結果與第六比較器的輸出結果進行邏輯與操作，並且該邏輯與操作的結果與第五比較器的輸出結果進行邏輯或操作以輸出第二控制信號，其中，第一過電流保護閾值電壓大於第二過電流保護閾值電壓。 According to another aspect of the present invention, an overcurrent protection circuit includes: a fourth comparison Comparing the second input voltage signal with the voltage control signal to output a high level when the voltage of the second input voltage signal is equal to the voltage of the voltage control signal; the fifth comparator, the second input voltage signal and the first The overcurrent protection threshold voltage is compared to output a high level when the voltage of the second input voltage signal is equal to the first overcurrent protection threshold voltage; the sixth comparator, the second input voltage signal and the second overcurrent protection threshold voltage Comparing, thereby outputting a high level when the voltage of the second input voltage signal is equal to the second overcurrent protection threshold voltage; wherein the output result of the fourth comparator is logically ANDed with the output result of the sixth comparator, and The result of the logic and operation is logically ORed with the output of the fifth comparator to output a second control signal, wherein the first overcurrent protection threshold voltage is greater than the second overcurrent protection threshold voltage.

根據本發明的另一方面，當採樣電壓小於第二參考電壓時，第一誤差放大器輸出的第一電壓信號為第三電壓值，其中，第三電壓值大於第一電壓值，第一過電流保護閾值電壓被設定為在第一電壓信號為第一電壓值時第二電阻上的電壓值，並且第二過電流保護閾值電壓小於第一過電流保護閾值電壓。 According to another aspect of the present invention, when the sampling voltage is less than the second reference voltage, the first voltage signal output by the first error amplifier is a third voltage value, wherein the third voltage value is greater than the first voltage value, the first overcurrent The protection threshold voltage is set to a voltage value on the second resistance when the first voltage signal is the first voltage value, and the second overcurrent protection threshold voltage is less than the first overcurrent protection threshold voltage.

根據本發明的另一方面，當採樣電壓小於第二參考電壓時，在第二輸入電壓信號的電壓小於第一過電流保護閾值電壓時，第二控制信號為低電平，而在第二輸入電壓信號的電壓等於第一過電流保護閾值電壓時，第二控制信號為高電平；當採樣電壓等於第二參考電壓時，第一誤差放大器輸出的第一電壓信號在第一電壓值和第二電壓值之間變化，當第二輸入電壓信號的電壓等於在電壓控制信號和第二過電流保護閾值電壓中的較高電壓與第一過電流保護閾值電壓中選擇的較低的電壓時，第二控制信號為高電平；當第二輸入電壓信號的電壓小於電壓控制信號和第二過電流保護閾值電壓中的至少一個並且小於第一過電流保護閾值電壓時，第二控制信號為低電平；當採樣電壓大於第二參考電壓並小於第三參考電壓時或者當採樣電壓等於第三參考電壓時，第一誤差放大器輸出的第一電壓信號為低電平，當第二輸入電壓信號的電壓小於第二過電流保護閾值電壓時， 第二控制信號為低電平，而當第二輸入電壓信號的電壓等於第二過電流保護閾值電壓時，第二控制信號為高電平。 According to another aspect of the present invention, when the sampling voltage is less than the second reference voltage, when the voltage of the second input voltage signal is less than the first overcurrent protection threshold voltage, the second control signal is at a low level, and at the second input When the voltage of the voltage signal is equal to the first overcurrent protection threshold voltage, the second control signal is at a high level; when the sampling voltage is equal to the second reference voltage, the first voltage signal output by the first error amplifier is at the first voltage value and Changing between two voltage values, when the voltage of the second input voltage signal is equal to a lower voltage selected between a higher voltage of the voltage control signal and the second overcurrent protection threshold voltage and a first overcurrent protection threshold voltage, The second control signal is at a high level; when the voltage of the second input voltage signal is less than at least one of the voltage control signal and the second overcurrent protection threshold voltage and is less than the first overcurrent protection threshold voltage, the second control signal is low Level; when the sampling voltage is greater than the second reference voltage and less than the third reference voltage or when the sampling voltage is equal to the third reference voltage, the first error A first amplifier output voltage signal is low, when the voltage of the second input voltage signal is less than a second over-current protection threshold voltage, The second control signal is at a low level, and when the voltage of the second input voltage signal is equal to the second overcurrent protection threshold voltage, the second control signal is at a high level.

根據本發明的另一方面，在第二控制信號為高電平時，邏輯驅動電路輸出低電平的第三控制信號，而在第二控制信號為低電平而第一控制信號為高電平時，邏輯驅動電路輸出高電平的第三控制信號。 According to another aspect of the present invention, when the second control signal is at a high level, the logic driving circuit outputs a third control signal of a low level, and when the second control signal is at a low level and the first control signal is at a high level The logic drive circuit outputs a third control signal of a high level.

根據本發明的另一方面，提供了一種包括如上所述的充電控制電路的反激式電源變換系統，所述反激式電源變換系統還包括：整流電路，將從交流電源輸入的電壓信號進行整流並輸入至充電電路；充電電路，基於整流電路輸入的電壓信號以及從充電控制電路輸出的第三控制信號來對電池進行充電，其中，在充電電路中，原邊繞組、開關、電流檢測電阻依次串聯連接，電流檢測電阻的另一端接地，電流檢測電阻上的電壓信號作為充電控制電路中的第二輸入電壓信號而輸入至充電控制電路，充電控制電路輸出的第三控制信號用於控制開關的導通和關斷；充電電路中的副邊繞組的輸出電壓用於對電池進行充電；充電電路中的輔助繞組反映副邊繞組輸出電壓的變化，並且連接有串聯連接的第三電阻和第四電阻，其中，第四電阻的另一端接地，第四電阻上的電壓信號作為充電控制電路的第一輸入電壓信號將副邊繞組對電池進行充電的輸出電壓回饋到充電控制電路。 According to another aspect of the present invention, there is provided a flyback power conversion system including a charge control circuit as described above, the flyback power conversion system further comprising: a rectifier circuit for performing a voltage signal input from an alternating current power source Rectifying and inputting to the charging circuit; charging circuit, charging the battery based on the voltage signal input by the rectifier circuit and the third control signal outputted from the charging control circuit, wherein, in the charging circuit, the primary winding, the switch, and the current detecting resistor Connected in series, the other end of the current detecting resistor is grounded, and the voltage signal on the current detecting resistor is input to the charging control circuit as a second input voltage signal in the charging control circuit, and the third control signal outputted by the charging control circuit is used to control the switch Turning on and off; the output voltage of the secondary winding in the charging circuit is used to charge the battery; the auxiliary winding in the charging circuit reflects the change of the output voltage of the secondary winding, and is connected with the third resistor and the fourth connected in series a resistor, wherein the other end of the fourth resistor is grounded, and the fourth resistor As the charge control voltage signal input circuit, a first voltage signal to the secondary winding to charge the battery output voltage feedback to the charge control circuit.

根據本發明的另一方面，在第一恒流充電控制模式下，副邊繞組以第一恒定電流對電池進行充電；在第一恒壓充電控制模式下，副邊繞組以第一恒定電壓對電池進行充電；在第二恒流充電控制模式下，副邊繞組以第二恒定電流對電池進行充電；在第二恒壓充電控制模式下，副邊繞組以第二恒定電壓對電池進行充電；其中，第一恒定電流大於第二恒定電流，第二恒定電壓大於第一恒定電壓。 According to another aspect of the present invention, in the first constant current charging control mode, the secondary winding charges the battery with a first constant current; in the first constant voltage charging control mode, the secondary winding has a first constant voltage pair The battery is charged; in the second constant current charging control mode, the secondary winding charges the battery with a second constant current; in the second constant voltage charging control mode, the secondary winding charges the battery with a second constant voltage; Wherein the first constant current is greater than the second constant current and the second constant voltage is greater than the first constant voltage.

根據本發明的另一方面，提供了一種在反激式電源變換系統中對電池進行充電的充電控制方法，包括：基於反映反激式電源變換系統 中的電池的充電電壓的變化的第一輸入電壓信號選擇第一恒流充電控制模式、第一恒壓充電控制模式、第二恒流充電控制模式以及第二恒壓充電控制模式中的一種充電控制模式，並產生與選擇的充電控制模式有關的第一控制信號以及電壓控制信號；將與反激式電源變換系統中的原邊繞組串聯連接的反應原邊繞組的電流變化的第二輸入電壓信號與電壓控制信號、第一過電流保護閾值電壓、第二過電流保護閾值電壓進行比較產生第二控制信號；基於第一控制信號和第二控制信號產生第三控制信號來控制反激式電源變換系統中與原邊繞組串聯連接的開關以對反激式電源變換系統的充電操作進行控制。 According to another aspect of the present invention, there is provided a charging control method for charging a battery in a flyback power conversion system, comprising: a reflection-based power conversion system based on reflection The first input voltage signal of the change of the charging voltage of the battery in the battery selects one of the first constant current charging control mode, the first constant voltage charging control mode, the second constant current charging control mode, and the second constant voltage charging control mode Controlling a mode and generating a first control signal associated with the selected charge control mode and a voltage control signal; a second input voltage that varies in current of the reactive primary winding connected in series with the primary winding in the flyback power conversion system The signal is compared with the voltage control signal, the first overcurrent protection threshold voltage, and the second overcurrent protection threshold voltage to generate a second control signal; and the third control signal is generated based on the first control signal and the second control signal to control the flyback power supply A switch in series with the primary winding in the conversion system controls the charging operation of the flyback power conversion system.

根據本發明的另一方面，當電池的充電電壓小於第一預定值時，選擇第一恒流充電控制模式從而以第一恒定電流對電池進行充電；在電池的充電電壓等於第一預定值時，選擇第一恒壓充電控制模式從而以第一恒定電壓對電池進行充電；在電池的充電電壓大於第一預定值而小於第二預定值時，選擇第二恒流充電控制模式從而以第二恒定電流對電池進行充電；在電池的充電電壓等於第二預定值時，選擇第二恒壓充電控制模式，從而以第二恒定電壓對電池進行充電；其中，第一恒定電流大於第二恒定電流，第二恒定電壓大於第一恒定電壓。 According to another aspect of the present invention, when the charging voltage of the battery is less than the first predetermined value, the first constant current charging control mode is selected to charge the battery with the first constant current; when the charging voltage of the battery is equal to the first predetermined value Selecting a first constant voltage charging control mode to charge the battery at a first constant voltage; selecting a second constant current charging control mode to be second when the charging voltage of the battery is greater than a first predetermined value and less than a second predetermined value The constant current charges the battery; when the charging voltage of the battery is equal to the second predetermined value, the second constant voltage charging control mode is selected to charge the battery at the second constant voltage; wherein the first constant current is greater than the second constant current The second constant voltage is greater than the first constant voltage.

根據本發明的另一方面，在第二控制信號為高電平時，產生低電平的第三控制信號從而使開關關斷；在第二控制信號為低電平而第一控制信號為高電平時，產生高電平的第三控制信號從而使開關導通。 According to another aspect of the present invention, when the second control signal is at a high level, a third control signal of a low level is generated to turn off the switch; when the second control signal is at a low level and the first control signal is high Normally, a third high level control signal is generated to turn the switch on.

110,510‧‧‧整流電路 110,510‧‧‧Rectifier circuit

120,520‧‧‧充電電路 120,520‧‧‧Charging circuit

130,530‧‧‧充電控制電路 130,530‧‧‧Charging control circuit

5310‧‧‧模式選擇電路 5310‧‧‧ mode selection circuit

5320‧‧‧邏輯驅動電路 5320‧‧‧Logical drive circuit

5330‧‧‧過電流保護電路 5330‧‧‧Overcurrent protection circuit

610‧‧‧採樣控制器 610‧‧‧Sampling controller

620‧‧‧退磁檢測器 620‧‧‧Demagnetization detector

630‧‧‧斜坡信號發生器 630‧‧‧Ramp signal generator

640‧‧‧第一恒流充電控制電路 640‧‧‧First constant current charging control circuit

650‧‧‧第二恒流充電控制電路 650‧‧‧Second constant current charging control circuit

680‧‧‧第一恒壓充電控制電路 680‧‧‧First constant voltage charging control circuit

660‧‧‧第二恒壓充電控制電路 660‧‧‧Second constant voltage charging control circuit

670‧‧‧控制電壓輸出電路 670‧‧‧Control voltage output circuit

6710‧‧‧緩衝器 6710‧‧‧buffer

6720‧‧‧低通濾波器 6720‧‧‧low pass filter

A1,A2,A3,A4‧‧‧及閘 A1, A2, A3, A4‧‧‧ and gate

AVDD‧‧‧電源電壓 AVDD‧‧‧Power supply voltage

C1,C2,C6,C9,C11,C20‧‧‧電容 C1, C2, C6, C9, C11, C20‧‧‧ capacitors

CMP0,CMP1,CMP2,CMP3,cmp_cc,cmp_ccl‧‧‧比較器 CMP0, CMP1, CMP2, CMP3, cmp_cc, cmp_ccl‧‧‧ comparator

C0‧‧‧電容器/電容 C0‧‧‧Capacitor/Capacitor

CV1,CV2,CC_high,CC_low‧‧‧電平信號 CV1, CV2, CC_high, CC_low‧‧‧ level signals

D1,D3‧‧‧二極體 D1, D3‧‧‧ diode

DRV‧‧‧端子 DRV‧‧‧ terminal

demag‧‧‧退磁信號 Demag‧‧‧ demagnetization signal

EA1,EA2‧‧‧誤差放大器 EA1, EA2‧‧‧ error amplifier

Fcv1,Fcv2,Fcc_high,Fcc_low,Fmin‧‧‧頻率 Fcv1, Fcv2, Fcc_high, Fcc_low, Fmin‧‧‧ frequency

Fo,Fcc_h,Fcc_l‧‧‧工作頻率 Fo, Fcc_h, Fcc_l‧‧‧ working frequency

G1,G2‧‧‧或閘 G1, G2‧‧‧ or gate

GD8‧‧‧驅動器 GD8‧‧‧ drive

I1,I1’,I2,I2’‧‧‧電流鏡 I1, I1', I2, I2'‧‧‧ current mirror

Io‧‧‧輸出電流/充電電流 Io‧‧‧Output current / charging current

Ipri‧‧‧輸出電流 Ipri‧‧‧Output current

Isec‧‧‧副邊輸出電流 Isec‧‧‧ secondary output current

Icc_l‧‧‧電流 Icc_l‧‧‧ Current

Ks‧‧‧採樣開關 Ks‧‧ sampling switch

LEB‧‧‧前沿消隱電路 LEB‧‧‧ leading edge blanking circuit

N9,N11‧‧‧反閘 N9, N11‧‧‧ reverse gate

Np‧‧‧原邊繞組 Np‧‧‧ primary winding

Nsec‧‧‧副邊繞組 Nsec‧‧‧ secondary winding

Naux‧‧‧輔助繞組 Naux‧‧‧Auxiliary winding

NG1,NG2‧‧‧反或閘 NG1, NG2‧‧‧ reverse or gate

Q,Q1,K1,K1’,K2,K2’‧‧‧開關 Q, Q1, K1, K1', K2, K2'‧‧‧ switch

R1,R2‧‧‧電阻 R1, R2‧‧‧ resistance

Rc1,Rc2‧‧‧分壓電阻 Rc1, Rc2‧‧‧ voltage resistor

Rs‧‧‧電流檢測電阻 Rs‧‧‧ current sense resistor

Req‧‧‧輸出線等效電阻 Req‧‧‧output line equivalent resistance

Sm-sw‧‧‧採樣控制信號 Sm-sw‧‧‧Sampling control signal

s1,s2‧‧‧控制信號 S1, s2‧‧‧ control signal

Sdry‧‧‧驅動控制信號 Sdry‧‧‧ drive control signal

Vcs‧‧‧電壓/電壓信號 Vcs‧‧‧voltage/voltage signal

V1,V2,V3,V4,Vaux,Ve,Vf,Vg,Vcs_peak‧‧‧電壓 V1, V2, V3, V4, Vaux, Ve, Vf, Vg, Vcs_peak‧‧‧ voltage

Vctrl‧‧‧電壓控制信號 Vctrl‧‧‧ voltage control signal

Vc0‧‧‧採樣電壓 Vc0‧‧‧Sampling voltage

V_FB‧‧‧回饋電壓 V _FB ‧‧‧ feedback voltage

VCO‧‧‧壓控振盪器 VCO‧‧‧voltage controlled oscillator

Vea1,Vea2,Vin,Vramp_cc,Vramp_ccl‧‧‧電壓信號 Vea1, Vea2, Vin, Vramp_cc, Vramp_ccl‧‧‧ voltage signal

Vref,Vref1,Vref2,Vd,Va‧‧‧參考電壓 Vref, Vref1, Vref2, Vd, Va‧‧‧ reference voltage

Vramp‧‧‧斜坡信號 Vramp‧‧‧ ramp signal

Vo‧‧‧充電電壓/輸出電壓 Vo‧‧‧Charging voltage/output voltage

Vth_max‧‧‧第一過電流保護閾值電壓 Vth_max‧‧‧First overcurrent protection threshold voltage

Vth_min‧‧‧第二過電流保護閾值電壓 Vth_min‧‧‧Second overcurrent protection threshold voltage

第1圖是根據現有技術的反激式電源變換系統的簡要示圖。 Fig. 1 is a schematic diagram of a flyback power conversion system according to the prior art.

第2圖示出了第1圖所示的反激式電源變換系統中回饋電壓、退磁信號、採 樣控制信號、輸出電流、原邊電流、以及電流檢測電阻上的電壓的波形變化時序圖。 Figure 2 shows the feedback voltage and demagnetization signal in the flyback power conversion system shown in Figure 1. A timing diagram of the waveform change of the control signal, the output current, the primary current, and the voltage across the current sense resistor.

第3圖和第4圖均示出了根據本發明示例性實施例的兩段式恒流加兩段式恒壓控制I-V曲線圖。 3 and 4 each show a two-stage constant current plus two-stage constant voltage control I-V graph according to an exemplary embodiment of the present invention.

第5圖示出了根據本發明示例性實施例的實現如第4圖所示的充電控制方式的反激式電源變換系統的框圖。 Fig. 5 is a block diagram showing a flyback power conversion system that implements a charge control mode as shown in Fig. 4, according to an exemplary embodiment of the present invention.

第6圖示出了根據本發明示例性實施例的充電控制電路中的模式選擇電路的示意圖。 Fig. 6 is a diagram showing a mode selection circuit in a charge control circuit according to an exemplary embodiment of the present invention.

第7圖示出了根據本發明示例性實施例的充電控制電路的過電流保護電路的示意圖。 FIG. 7 is a schematic diagram showing an overcurrent protection circuit of a charge control circuit according to an exemplary embodiment of the present invention.

第8圖示出了根據本發明示例性實施例的充電控制電路的邏輯驅動電路的電路示意圖。 FIG. 8 is a circuit diagram showing a logic driving circuit of a charging control circuit according to an exemplary embodiment of the present invention.

第9圖是根據本發明示例性實施例的第一恒流充電控制電路的示例性電路圖。 FIG. 9 is an exemplary circuit diagram of a first constant current charge control circuit according to an exemplary embodiment of the present invention.

第10圖是根據本發明示例性實施例的第一恒流充電控制模式下的時序圖。 FIG. 10 is a timing chart in a first constant current charging control mode according to an exemplary embodiment of the present invention.

第11圖是根據本發明示例性實施例的第二恒流充電控制電路的示例性電路圖。 Figure 11 is an exemplary circuit diagram of a second constant current charging control circuit in accordance with an exemplary embodiment of the present invention.

第12圖是根據本發明示例性實施例的第二恒壓充電控制電路的示例性電路圖。 Fig. 12 is an exemplary circuit diagram of a second constant voltage charge control circuit according to an exemplary embodiment of the present invention.

第13圖示出了壓控振盪器VCO的輸出頻率與輸入電壓之間的關係示圖。 Figure 13 shows a diagram showing the relationship between the output frequency of the voltage controlled oscillator VCO and the input voltage.

第14圖示出了根據本發明示例性實施例的根據電池的充電電壓的變化而改變的充電電池的充電電流、過電流保護的電壓以及開關的工作頻率Fo的變化的示意圖。 Fig. 14 is a view showing a change of a charging current of a rechargeable battery, a voltage of an overcurrent protection, and a change in an operating frequency Fo of a switch, which are changed according to a change in a charging voltage of a battery, according to an exemplary embodiment of the present invention.

下面將結合具體的實施例來對本發明進行詳細的描述。本領域技術人員應該理解，本發明所示的實施例只是示例性的，並不作為對本發明的限制。 The invention will now be described in detail in connection with the specific embodiments. Those skilled in the art should understand that the embodiments of the present invention are only exemplary and not intended to limit the invention.

第3圖和第4圖均示出了根據本發明示例性實施例的兩段式恒流加兩段式恒壓控制I-V曲線圖，其不同之處在於橫縱坐標的不同，從而使本領域技術人員能夠更清楚更直觀地瞭解根據本發明示例性實施例的採用兩段式恒流加兩段式恒壓控制方式的輸出電流Io和輸出電壓Vo的變化。根據第3圖和第4圖所示，當電池的充電電壓Vo小於電壓V1時，由與第一恒流充電控制模式相應的恒定的大電流Icc_h對電池進行快速充電；當電池電壓達到電壓V1時，則用與第一恒壓充電控制模式相應的恒定的電壓V1對電池進行充電，同時電池的充電電流逐漸減小；當電池的充電電流減小到Icc_l(Icc_l小於Icc_h)時，則進入第二恒流充電控制模式，使電池的充電電流維持在Icc_l；當電池電壓達到電壓V2(V2大於V1)時，則進入第二恒壓充電控制模式，從而以恒定電壓V2對電池進行充電，此時電池的充電電流由Icc_l繼續減小，直到為“0”，即不對電池充電。 3 and 4 each show a two-stage constant current plus two-stage constant pressure control IV graph according to an exemplary embodiment of the present invention, which differs in the difference in the horizontal and vertical coordinates, thereby enabling those skilled in the art to The change of the output current Io and the output voltage Vo using the two-stage constant current plus two-stage constant voltage control method according to an exemplary embodiment of the present invention is clearly understood more intuitively. According to FIGS. 3 and 4, when the charging voltage Vo of the battery is less than the voltage V1, the battery is rapidly charged by the constant large current Icc_h corresponding to the first constant current charging control mode; when the battery voltage reaches the voltage V1 At the same time, the battery is charged with a constant voltage V1 corresponding to the first constant voltage charging control mode, and the charging current of the battery is gradually decreased; when the charging current of the battery is reduced to Icc_l (Icc_l is less than Icc_h), then the battery enters The second constant current charging control mode maintains the charging current of the battery at Icc_l; when the battery voltage reaches the voltage V2 (V2 is greater than V1), the second constant voltage charging control mode is entered, thereby charging the battery with the constant voltage V2. At this time, the charging current of the battery continues to decrease by Icc_l until it is "0", that is, the battery is not charged.

第5圖示出了根據本發明示例性實施例的實現如第4圖所示的充電控制方式的反激式電源變換系統的框圖。 Fig. 5 is a block diagram showing a flyback power conversion system that implements a charge control mode as shown in Fig. 4, according to an exemplary embodiment of the present invention.

如第5圖所示，根據本發明示例性實施例的反激式電源變換系統包括整流電路510、充電電路520和充電控制電路530。其中，整流電路510對輸入的交流電源進行橋式整流並將橋式整流後所得的電壓信號Vin輸入到充電電路520。充電電路520包括原邊繞組Np、副邊繞組Nsec以及輔助繞組Naux以及與其各自相連的二極體、電阻以及電容等構成的輔助電路。原邊繞組連接到開關Q。在本發明中開關Q是以雙極性接面電晶體(Bipolar Junction Transistor,BJT)為例進行的描述，但是本領域技術人員應該理解，這裡的開關Q也可以為金屬氧化物半導體場效應管(Metal-Oxide-Semiconductor Field-Effect Transistor,MOSFET)以及絕緣柵 雙極性接面電晶體(Insulated Gate Bipolar Transistor,IGBT)等等開關電晶體。 As shown in FIG. 5, a flyback power conversion system according to an exemplary embodiment of the present invention includes a rectifier circuit 510, a charging circuit 520, and a charging control circuit 530. The rectifier circuit 510 performs bridge rectification on the input AC power source and inputs the voltage signal Vin obtained by the bridge rectification to the charging circuit 520. The charging circuit 520 includes an auxiliary winding circuit including a primary winding Np, a secondary winding Nsec, and an auxiliary winding Naux, and a diode, a resistor, and a capacitor connected thereto. The primary winding is connected to switch Q. In the present invention, the switch Q is described by taking a Bipolar Junction Transistor (BJT) as an example, but those skilled in the art should understand that the switch Q herein may also be a metal oxide semiconductor field effect transistor ( Metal-Oxide-Semiconductor Field-Effect Transistor (MOSFET) and insulated gate Switching transistor such as Insulated Gate Bipolar Transistor (IGBT).

這裡，根據本發明示例性實施例的整流電路510和充電電路520與現有技術中如第1圖所示的整流電路110以及充電電路120具有相同的結構。根據本發明示例性實施例的充電控制電路530包括模式選擇電路5310、邏輯驅動電路5320和過電流保護電路5330。本領域技術人員應該理解，這裡所述的充電控制電路530是相對獨立的控制晶片，其可以應用於反激式電源變換系統，也可以應用到其他電路系統而進行相應的控制。為了便於本領域技術人員更好地理解本發明，在本發明中將該充電控制電路530應用於反激式電源變換系統中以對該反激式電源變換系統中的開關Q進行控制，進而以兩段式恒流兩段式恒壓的控制方式對電池進行充電。 Here, the rectifying circuit 510 and the charging circuit 520 according to an exemplary embodiment of the present invention have the same configuration as the rectifying circuit 110 and the charging circuit 120 as shown in FIG. 1 in the prior art. The charge control circuit 530 according to an exemplary embodiment of the present invention includes a mode selection circuit 5310, a logic drive circuit 5320, and an overcurrent protection circuit 5330. Those skilled in the art will appreciate that the charge control circuit 530 described herein is a relatively independent control chip that can be applied to a flyback power conversion system or to other circuit systems for corresponding control. In order to facilitate the understanding of the present invention by those skilled in the art, in the present invention, the charging control circuit 530 is applied to a flyback power conversion system to control the switch Q in the flyback power conversion system, thereby The two-stage constant current two-stage constant voltage control method charges the battery.

這裡，模式選擇電路5310接收從FB處回饋的回饋電壓V_FB。模式選擇電路5310根據電池充電電壓Vo的變化選擇如第4圖所示的不同的控制模式。具體來講，模式選擇電路5310基於與電池的充電電壓Vo相應的從FB處回饋的回饋電壓V_FB選擇如第4圖所示的第一恒流充電控制模式、第一恒壓充電控制模式、第二恒流充電控制模式以及第二恒壓充電控制模式中的一種充電控制模式，並向邏輯驅動電路5320輸出與選擇的充電控制模式有關的控制信號s1，同時向過電流保護電路5330輸出電壓控制信號Vctrl。這裡，回饋電壓V_FB與電池的充電電壓Vo之間的關係如公式(1)所示，並且第一恒流充電控制模式下對電池進行充電的電流大於第二恒流充電控制模式下對電池進行充電的電流，而第一恒壓充電控制模式下的電池的充電電壓小於第二恒壓充電控制模式下電池的充電電壓。 Here, the mode selection circuit 5310 receives the feedback voltage V _FB fed back from the _FB . The mode selection circuit 5310 selects different control modes as shown in FIG. 4 in accordance with changes in the battery charging voltage Vo. Specifically, the mode selection circuit 5310 selects the first constant current charging control mode, the first constant voltage charging control mode, and the first constant voltage charging control mode as shown in FIG. 4 based on the feedback voltage V _FB fed back from the FB corresponding to the charging voltage Vo of the battery. a second constant current charging control mode and one of the second constant voltage charging control modes, and outputting a control signal s1 related to the selected charging control mode to the logic driving circuit 5320 while outputting a voltage to the overcurrent protection circuit 5330 Control signal Vctrl. Here, the relationship between the feedback voltage V _FB and the charging voltage Vo of the battery is as shown in the formula (1), and the current for charging the battery in the first constant current charging control mode is greater than the battery in the second constant current charging control mode. The charging current is performed, and the charging voltage of the battery in the first constant voltage charging control mode is smaller than the charging voltage of the battery in the second constant voltage charging control mode.

過電流保護電路5330除了接收從模式選擇電路5310輸出的電壓控制信號Vctrl之外，還接收通過開關Q與原邊繞組Np連接的電流檢測電阻Rs上的電壓信號Vcs(也即端子CS處的電壓信號)以及第一過電流保護閾值電壓Vth_max以及第二過電流保護閾值電壓Vth_min，其中，第一過電 流保護閾值電壓Vth_max大於第二過電流保護閾值電壓Vth_min。過電流保護電路5330將輸入的電壓控制信號Vctrl、第一過電流保護閾值電壓Vth_max以及第二過電流保護閾值電壓Vth_min分別與電流檢測電阻Rs上的電壓信號Vcs進行比較並向邏輯驅動電路5320輸出控制信號s2。 The overcurrent protection circuit 5330 receives the voltage signal Vcs (that is, the voltage at the terminal CS) on the current detecting resistor Rs connected to the primary winding Np through the switch Q in addition to the voltage control signal Vctrl outputted from the mode selection circuit 5310. a signal) and a first overcurrent protection threshold voltage Vth_max and a second overcurrent protection threshold voltage Vth_min, wherein the first overcurrent The flow protection threshold voltage Vth_max is greater than the second overcurrent protection threshold voltage Vth_min. The overcurrent protection circuit 5330 compares the input voltage control signal Vctrl, the first overcurrent protection threshold voltage Vth_max, and the second overcurrent protection threshold voltage Vth_min with the voltage signal Vcs on the current detecting resistor Rs and outputs it to the logic driving circuit 5320, respectively. Control signal s2.

當電池的充電電壓Vo小於電壓V1值並且電流檢測電阻Rs的電壓信號等於第一過電流保護閾值電壓Vth_max時，過電流保護電路5330輸出的控制信號s2為高電平。 When the charging voltage Vo of the battery is less than the voltage V1 value and the voltage signal of the current detecting resistor Rs is equal to the first overcurrent protection threshold voltage Vth_max, the control signal s2 output from the overcurrent protection circuit 5330 is at a high level.

當電池的充電電壓Vo等於電壓V1時，電壓控制信號Vctrl和第二過電流保護閾值電壓Vth_min中比較高的電壓和第一過電流保護閾值電壓Vth_max比較，選擇一個比較低的電壓，當電流檢測電阻Rs上的電壓信號Vcs等於該較低的電壓時，控制信號s2為高電平；而另外在電流檢測電阻Rs上的電壓信號Vcs小於電壓控制信號Vctrl和第二過電流保護閾值電壓Vth_min中的至少一個並且小於第一過電流保護閾值電壓Vth_max時，控制信號s2為低電平。 When the charging voltage Vo of the battery is equal to the voltage V1, the relatively high voltage of the voltage control signal Vctrl and the second overcurrent protection threshold voltage Vth_min is compared with the first overcurrent protection threshold voltage Vth_max, and a relatively low voltage is selected when the current is detected. When the voltage signal Vcs on the resistor Rs is equal to the lower voltage, the control signal s2 is at a high level; and the voltage signal Vcs on the current detecting resistor Rs is smaller than the voltage control signal Vctrl and the second overcurrent protection threshold voltage Vth_min. When at least one and less than the first overcurrent protection threshold voltage Vth_max, the control signal s2 is at a low level.

當電池的充電電壓Vo大於電壓V1小於電壓V2或者等於電壓V2時，在電流檢測電阻Rs上的電壓信號Vcs的電壓值等於第二過電流保護閾值電壓Vth_min時，控制信號s2為高電平信號。需要注意的是，無論在什麼情況下，當控制信號s2輸出高電平信號時，開關Q關斷。 When the charging voltage Vo of the battery is greater than the voltage V1 being less than the voltage V2 or equal to the voltage V2, when the voltage value of the voltage signal Vcs on the current detecting resistor Rs is equal to the second overcurrent protection threshold voltage Vth_min, the control signal s2 is a high level signal. . It should be noted that, in any case, when the control signal s2 outputs a high level signal, the switch Q is turned off.

邏輯驅動電路5320根據從模式選擇電路5310輸出的控制信號s1和過電流保護電路5330輸出的控制信號s2來輸出驅動控制信號Sdrv以對開關Q的導通和關斷進行控制。作為一種示例，驅動控制信號Sdrv通過端子DRV輸出到開關Q以對其進行控制，進而對反激式電源變換系統進行控制。 The logic drive circuit 5320 outputs a drive control signal Sdrv based on the control signal s1 output from the mode selection circuit 5310 and the control signal s2 output from the overcurrent protection circuit 5330 to control the on and off of the switch Q. As an example, the drive control signal Sdrv is output to the switch Q through the terminal DRV to control it, thereby controlling the flyback power conversion system.

根據第5圖所示的反激式電源變換系統，可以以兩段恒壓加兩段恒流的控制方式對電池進行充電，從而實現了電池快速充電並優化了電池壽命。 According to the flyback power conversion system shown in FIG. 5, the battery can be charged by two constant voltage plus two constant current control modes, thereby realizing rapid battery charging and optimizing battery life.

第6圖示出了根據本發明示例性實施例的充電控制電路530中的模式選擇電路5310的示意圖。本領域技術人員應該理解，第6圖所示的電路只是示例性的，並不作為對本發明的限制。 FIG. 6 shows a schematic diagram of mode selection circuit 5310 in charge control circuit 530, in accordance with an exemplary embodiment of the present invention. It should be understood by those skilled in the art that the circuit shown in FIG. 6 is merely exemplary and is not intended to limit the invention.

根據第6圖所示，所述模式選擇電路5310包括退磁檢測器620、第一恒流充電控制電路640、第一恒壓充電控制電路680、第二恒流充電控制電路650、第二恒壓充電控制電路660以及控制電壓輸出電路670。其中，第一恒壓充電控制電路680包括採樣控制器610、斜坡信號發生器630、誤差放大器EA1、比較器CMP0以及採樣開關Ks和電容器C0。 According to FIG. 6, the mode selection circuit 5310 includes a demagnetization detector 620, a first constant current charging control circuit 640, a first constant voltage charging control circuit 680, a second constant current charging control circuit 650, and a second constant voltage. The charge control circuit 660 and the control voltage output circuit 670. The first constant voltage charging control circuit 680 includes a sampling controller 610, a ramp signal generator 630, an error amplifier EA1, a comparator CMP0, and a sampling switch Ks and a capacitor C0.

如前所述，在開關Q導通時，變壓器儲存能量，變壓器的原邊繞組Np的電流線性上升，電流檢測電阻Rs上的電壓Vcs(作為一種示例，也可以稱作端子CS處的電壓)也線性上升。在開關Q關斷期間，變壓器上儲存的能量釋放到輸出端，退磁開始，此時輔助繞組Naux的電壓Vaux映射副邊繞組Nsec的輸出電壓，也即電池的充電電壓，因為輔助繞組Naux的電阻R2上的回饋電壓V_FB為輔助繞組Naux的輸出電壓Vaux的分壓，也即所述回饋電壓V_FB也可反映電池的充電電壓的大小，因此可通過比較器比較回饋電壓V_FB高於某個參考電壓Vd(例如0.1V)來判斷退磁的開始和結束。 As described above, when the switch Q is turned on, the transformer stores energy, the current of the primary winding Np of the transformer rises linearly, and the voltage Vcs on the current detecting resistor Rs (which may also be referred to as voltage at the terminal CS as an example) Linear rise. During the switch Q off, the energy stored on the transformer is released to the output, and demagnetization begins. At this time, the voltage Vaux of the auxiliary winding Naux maps the output voltage of the secondary winding Nsec, that is, the charging voltage of the battery, because the resistance of the auxiliary winding Naux The feedback voltage V _FB on R2 is the divided voltage of the output voltage Vaux of the auxiliary winding Naux, that is, the feedback voltage V _FB can also reflect the magnitude of the charging voltage of the battery, so the comparator can compare the feedback voltage V _FB higher than a certain The reference voltage Vd (for example, 0.1 V) is used to judge the start and end of demagnetization.

這裡，回饋電壓V_FB輸入到退磁檢測器620。在退磁檢測器620中設置有比較器以將回饋電壓V_FB與參考電壓Vd進行比較，並輸出退磁信號demag。當退磁檢測器620中設置的比較器判斷出回饋電壓V_FB大於該參考電壓Vd時，輸出高電平的退磁信號demag，而當回饋電壓V_FB小於該參考電壓Vd時，輸出低電平的退磁信號demag。也即，在副邊繞組Nsec處於退磁區間時，退磁信號demag為高電平，否則，退磁信號demag為低電平。退磁檢測器620向第一恒壓充電控制電路680、第一恒流充電控制電路640和第二恒流充電控制電路650輸出退磁信號demag，而上述部件根據從退磁檢測器620接收的退磁信號demag進行相應的操作。 Here, the feedback voltage V _{FB is} input to the demagnetization detector 620. A comparator is provided in the demagnetization detector 620 to compare the feedback voltage V _FB with the reference voltage Vd and output a demagnetization signal demag. When the comparator set in the demagnetization detector 620 determines that the feedback voltage V _{FB is} greater than the reference voltage Vd, the demagnetization signal demag of the high level is output, and when the feedback voltage V _{FB is} less than the reference voltage Vd, the low level is output. Demagnetization signal demag. That is, the demagnetization signal demag is at a high level when the secondary winding Nsec is in the demagnetization section, and otherwise, the demagnetization signal demag is at a low level. The demagnetization detector 620 outputs a demagnetization signal demag to the first constant voltage charge control circuit 680, the first constant current charge control circuit 640, and the second constant current charge control circuit 650, and the above components are based on the demagnetization signal demag received from the demagnetization detector 620. Take the appropriate action.

退磁檢測器620的退磁信號demag輸入至第一恒壓充電控制 電路680中的採樣控制器610和斜坡信號發生器630。採樣控制器610根據從退磁檢測器620輸出的退磁信號demag生成採樣控制信號Sm_sw以控制採樣開關Ks，而斜坡信號發生器630根據退磁信號demag生成斜坡信號Vramp。 The demagnetization signal demag of the demagnetization detector 620 is input to the first constant voltage charging control Sample controller 610 and ramp signal generator 630 in circuit 680. The sampling controller 610 generates a sampling control signal Sm_sw based on the demagnetization signal demag output from the demagnetization detector 620 to control the sampling switch Ks, and the ramp signal generator 630 generates a ramp signal Vramp based on the demagnetization signal demag.

具體來講，採樣控制器610根據退磁檢測器620輸出的退磁信號demag來生成採樣控制信號Sm_sw，所述採樣控制信號Sm_sw用於控制採樣開關Ks的通斷。在退磁開始時，採樣控制器610根據退磁信號demag生成高電平的採樣控制信號Sm_sw，使得採樣開關Ks閉合，從而回饋電壓V_FB輸入至誤差放大器EA1。在退磁過程幾乎結束，也即流經副邊繞組Nsec的副邊輸出電流Isec接近零(例如第2圖中所示的A點)時，採樣控制信號Sm_sw由高電平轉為低電平，採樣開關Ks回應採樣控制信號Sm_sw由高電平到低電平的轉變而斷開，此刻的回饋電壓V_FB經由一端連接至採樣開關Ks一端接地的電容C0被保持為誤差放大器EA1的輸入電壓信號。為了簡便起見，可以將採樣開關Ks閉合時輸入到誤差放大器EA1的回饋電壓V_FB或者採樣開關Ks斷開時所獲得的回饋電壓V_FB統稱為採樣電壓Vc0。第一恒壓充電控制電路680中的誤差放大器EA1將採樣電壓Vc0與參考電壓Vref1(例如2V)進行比較，並將採樣電壓Vc0與參考電壓Vref1之間的差值放大並輸出電壓信號Vea1。這裡，所述採樣電壓Vc0輸入到誤差放大器EA1的負向輸入端，而參考電壓Vref1輸入至所述誤差放大器EA1的正向輸入端。這裡誤差放大器EA1的輸出端分別與控制電壓輸出電路670和第一恒壓充電控制電路680中的比較器CMP0連接，以將電壓信號Vea1輸出到上述這兩個元件。 Specifically, the sampling controller 610 generates a sampling control signal Sm_sw according to the demagnetization signal demag output from the demagnetization detector 620, and the sampling control signal Sm_sw is used to control the on/off of the sampling switch Ks. At the start of demagnetization, the sampling controller 610 generates a high-level sampling control signal Sm_sw according to the demagnetization signal demag, so that the sampling switch Ks is closed, so that the feedback voltage V _{FB is} input to the error amplifier EA1. When the demagnetization process is almost finished, that is, the secondary side output current Isec flowing through the secondary winding Nsec is close to zero (for example, point A shown in FIG. 2), the sampling control signal Sm_sw is changed from the high level to the low level. The sampling switch Ks is turned off in response to the transition of the sampling control signal Sm_sw from a high level to a low level. The feedback voltage V _FB at this moment is held as an input voltage signal of the error amplifier EA1 via a capacitor C0 whose one end is connected to the ground of the sampling switch Ks. . Input sake of simplicity, the sampling switch Ks may be closed to the feedback voltage V _FB or sampling error amplifier EA1 OFF switch Ks obtained referred to as the feedback voltage V _FB sampled voltage Vc0. The error amplifier EA1 in the first constant voltage charging control circuit 680 compares the sampling voltage Vc0 with the reference voltage Vref1 (for example, 2V), and amplifies the difference between the sampling voltage Vc0 and the reference voltage Vref1 and outputs the voltage signal Vea1. Here, the sampling voltage Vc0 is input to the negative input terminal of the error amplifier EA1, and the reference voltage Vref1 is input to the forward input terminal of the error amplifier EA1. Here, the output of the error amplifier EA1 is connected to the control voltage output circuit 670 and the comparator CMP0 in the first constant voltage charge control circuit 680, respectively, to output the voltage signal Vea1 to the above two elements.

退磁檢測器620輸出的退磁信號demag還輸入到第一恒壓充電控制電路680中的斜坡信號發生器630以對斜坡信號發生器630進行控制。具體來講，當退磁信號demag為高電平時，控制斜坡信號發生器630的重定開關以對斜坡信號Vramp進行重定。這裡，斜坡信號Vramp的大小在V3-V4之間變化。具體來講，所述斜坡信號發生器630在退磁信號demag變為高電平時，將斜坡信號Vramp的電壓重定到電壓V4，斜坡信號Vramp從電 壓V4在下降時間內逐漸降低到電壓V3(這裡，V4>V3)。所述的逐漸降低表示單調連續地降低，可以是線性降低，也可以是非線性降低。 The demagnetization signal demag output from the demagnetization detector 620 is also input to the ramp signal generator 630 in the first constant voltage charge control circuit 680 to control the ramp signal generator 630. Specifically, when the demagnetization signal demag is at a high level, the reset switch of the ramp signal generator 630 is controlled to reset the ramp signal Vramp. Here, the magnitude of the ramp signal Vramp varies between V3-V4. Specifically, the ramp signal generator 630 resets the voltage of the ramp signal Vramp to the voltage V4 when the demagnetization signal demag becomes a high level, and the ramp signal Vramp is charged. The voltage V4 gradually decreases to a voltage V3 during the fall time (here, V4 > V3). The gradual decrease indicates a monotonous continuous decrease, which may be a linear decrease or a non-linear decrease.

從誤差放大器EA1輸出的電壓信號Vea1輸入到比較器CMP0的正向輸入端，而從斜坡信號發生器630輸出的斜坡信號Vramp輸入到比較器CMP0的反向輸入端，從而比較器CMP0可以將輸入的電壓信號Vea1和斜坡信號Vramp進行比較並輸出電平信號CV1，當電池的充電電壓Vo小於電壓V1時(也即採樣電壓Vc0小於參考電壓Vref1時)，電平信號CV1輸出為高電平，而當電池的充電電壓Vo等於電壓V1時(也即採樣電壓Vc0等於參考電壓Vref1時)，電平信號CV1為具有第一頻率的在高電平和低電平之間變換的電平信號。 The voltage signal Vea1 output from the error amplifier EA1 is input to the forward input terminal of the comparator CMP0, and the ramp signal Vramp output from the ramp signal generator 630 is input to the inverting input terminal of the comparator CMP0, so that the comparator CMP0 can input The voltage signal Vea1 is compared with the ramp signal Vramp and outputs a level signal CV1. When the charging voltage Vo of the battery is less than the voltage V1 (that is, when the sampling voltage Vc0 is smaller than the reference voltage Vref1), the level signal CV1 is outputted to a high level. When the charging voltage Vo of the battery is equal to the voltage V1 (that is, when the sampling voltage Vc0 is equal to the reference voltage Vref1), the level signal CV1 is a level signal having a first frequency that is converted between a high level and a low level.

控制電壓輸出電路670接收從第一恒壓充電控制電路680輸入的電壓信號Vea1，並且該控制電壓輸出電路670包括兩個串聯的分壓電阻Rc1和Rc2和低通濾波器6720。其中，低通濾波器6720連接在兩個分壓電阻的連接點處以對分壓電阻Rc2上的電壓信號進行低通濾波並輸出分壓電阻Rc2上的電壓信號，也即電壓控制信號Vctrl。 The control voltage output circuit 670 receives the voltage signal Vea1 input from the first constant voltage charge control circuit 680, and the control voltage output circuit 670 includes two series-divided voltage dividing resistors Rc1 and Rc2 and a low-pass filter 6720. The low-pass filter 6720 is connected to the connection point of the two voltage dividing resistors to low-pass filter the voltage signal on the voltage dividing resistor Rc2 and output the voltage signal on the voltage dividing resistor Rc2, that is, the voltage control signal Vctrl.

作為另一實施例，所述控制電壓輸出電路670另外還可包括緩衝器6710，所述緩衝器6710與分壓電阻Rc1和Rc2串聯，用於對從誤差放大器EA1輸出的電壓信號Vea1進行緩衝並將緩衝後的電壓信號輸入到所述兩個分壓電阻Rc1和Rc2，以增強電壓信號Vea1的驅動能力。 As another embodiment, the control voltage output circuit 670 may further include a buffer 6710 connected in series with the voltage dividing resistors Rc1 and Rc2 for buffering the voltage signal Vea1 output from the error amplifier EA1. The buffered voltage signal is input to the two voltage dividing resistors Rc1 and Rc2 to enhance the driving ability of the voltage signal Vea1.

作為又一實施例，所述控制電壓輸出電路670另外還可以包括電容C6，其可以與分壓電阻Rc1和Rc2構成的串聯電路(在此情況下，所述控制電壓輸出電路670可不包括緩衝器6710)並聯或者與緩衝器6710、分壓電阻Rc1和Rc2共同形成的串聯電路並聯，以使得回饋環路更加穩定。 As still another embodiment, the control voltage output circuit 670 may additionally include a capacitor C6, which may be in series with the voltage dividing resistors Rc1 and Rc2 (in this case, the control voltage output circuit 670 may not include a buffer 6710) Parallel or in parallel with the series circuit formed by the buffer 6710 and the voltage dividing resistors Rc1 and Rc2 to make the feedback loop more stable.

對於本領域技術人員來說，需要明確的是，所述控制電壓輸出電路670可以包括所述緩衝器6710和電容C6中的至少一個，甚至可以不包括所述緩衝器6710和電容C6。 It will be apparent to those skilled in the art that the control voltage output circuit 670 may include at least one of the buffer 6710 and the capacitor C6, and may not even include the buffer 6710 and the capacitor C6.

第一恒流充電控制電路640根據從退磁檢測器620輸出的退磁信號demag生成具有第二頻率的其電壓值在高電平和低電平之間變換的電平信號CC_high。第二恒流充電控制電路650根據從退磁檢測器620輸出的退磁信號demag生成具有第三頻率的其電壓值在高電平和低電平之間變換的電平信號CC_low。其中，所述第二頻率高於所述第三頻率。 The first constant current charging control circuit 640 generates a level signal CC_high having a second frequency whose voltage value is converted between a high level and a low level based on the demagnetization signal demag output from the demagnetization detector 620. The second constant current charging control circuit 650 generates a level signal CC_low having a third frequency whose value is converted between a high level and a low level in accordance with the demagnetization signal demag output from the demagnetization detector 620. Wherein the second frequency is higher than the third frequency.

第二恒壓充電控制電路660將電容C0上的電壓(也即採樣電壓Vc0)與參考電壓Vref2進行比較並輸出電平信號CV2，其中電平信號CV2在對電池充電的充電電壓Vo小於電壓V2(即採樣電壓Vc0小於參考電壓Vref2)時，電平信號CV2具有高電平；而在對電池充電的充電電壓Vo等於電壓V2(即採樣電壓Vc0等於參考電壓Vref2)時，輸出具有第四頻率的其電壓值在高電平和低電平之間變換的電平信號CV2。其中，第四頻率低於第二頻率和第三頻率。 The second constant voltage charging control circuit 660 compares the voltage on the capacitor C0 (that is, the sampling voltage Vc0) with the reference voltage Vref2 and outputs a level signal CV2, wherein the charging signal Vo for charging the battery to the level signal CV2 is smaller than the voltage V2. (ie, when the sampling voltage Vc0 is smaller than the reference voltage Vref2), the level signal CV2 has a high level; and when the charging voltage Vo for charging the battery is equal to the voltage V2 (ie, the sampling voltage Vc0 is equal to the reference voltage Vref2), the output has the fourth frequency. The level signal CV2 whose voltage value is converted between a high level and a low level. Wherein the fourth frequency is lower than the second frequency and the third frequency.

根據本發明示例性實施例，從第一恒流充電控制電路640輸出的電平信號CC_high與從第二恒壓充電控制電路660輸出的電平信號CV2輸入至及閘A1進行邏輯與操作，及閘A1的輸出信號與電平信號CV1輸入至及閘A2進行邏輯與操作。從第二恒流充電控制電路650輸出的電平信號CC_low與從第二恒壓充電控制電路660輸出的電平信號CV2輸入至及閘A3進行邏輯與操作，並且及閘A3的輸出信號和及閘A2的輸出信號被輸入至或閘G1進行邏輯或操作，並從其輸出控制信號s1。稍後將對所述四種充電控制模式的工作原理以及示例性的電路結構進行詳細的介紹。 According to an exemplary embodiment of the present invention, the level signal CC_high outputted from the first constant current charging control circuit 640 and the level signal CV2 outputted from the second constant voltage charging control circuit 660 are input to the AND gate A1 for logical AND operation, and The output signal of the gate A1 and the level signal CV1 are input to the AND gate A2 for logical AND operation. The level signal CC_low outputted from the second constant current charging control circuit 650 and the level signal CV2 outputted from the second constant voltage charging control circuit 660 are input to the AND gate A3 for logical AND operation, and the output signal of the gate A3 and The output signal of the gate A2 is input to the OR gate G1 for logical OR operation, and the control signal s1 is output therefrom. The working principle of the four charging control modes and an exemplary circuit configuration will be described in detail later.

第7圖示出了根據本發明示例性實施例的充電控制電路530的過電流保護電路5330的示意圖。所述過電流保護電路5330包括三個比較器CMP1,CMP2和CMP3，一個及閘A4、一個或閘G2以及前沿消隱電路LEB。在變壓器原邊繞組側通過開關Q與原邊繞組Np相連的電流檢測電阻Rs上的電壓信號Vcs經用於消除開關Q每次導通所產生的干擾的前沿消隱電路LEB分別被輸入到比較器CMP1,CMP2以及CMP3的正向輸入端，比較器 CMP1,CMP2和CMP3的反向輸入端分別輸入了模式選擇電路5310輸出的電壓控制信號Vctrl、第二過電流保護閾值電壓Vth_min和第一過電流保護閾值電壓Vth_max，以分別與電流檢測電阻Rs上的電壓信號Vcs進行比較。其中，比較器CMP1和CMP2輸出的比較結果作為及閘A4的輸入信號以進行邏輯與操作，並將進行邏輯與操作之後的輸出信號輸入到或閘G2的一個輸入端。或閘G2的另一個輸入端接收來自比較器CMP3的輸出信號。 FIG. 7 shows a schematic diagram of an overcurrent protection circuit 5330 of the charge control circuit 530 according to an exemplary embodiment of the present invention. The overcurrent protection circuit 5330 includes three comparators CMP1, CMP2 and CMP3, a sum gate A4, an OR gate G2, and a leading edge blanking circuit LEB. The voltage signal Vcs on the current detecting resistor Rs connected to the primary winding Np through the switch Q on the primary winding side of the transformer is input to the comparator through the leading edge blanking circuit LEB for eliminating the interference generated by the switch Q being turned on each time. Positive input of CMP1, CMP2 and CMP3, comparator The inverting input terminals of CMP1, CMP2 and CMP3 respectively input the voltage control signal Vctrl outputted by the mode selection circuit 5310, the second overcurrent protection threshold voltage Vth_min and the first overcurrent protection threshold voltage Vth_max to respectively correspond to the current detecting resistor Rs. The voltage signal Vcs is compared. The comparison result of the outputs of the comparators CMP1 and CMP2 is used as an input signal of the AND gate A4 to perform a logical AND operation, and an output signal after the logical AND operation is input to an input terminal of the gate G2. The other input of OR gate G2 receives the output signal from comparator CMP3.

根據第7圖所示的電路，當電流檢測電阻上的電壓Vcs達到第一過電流保護閾值電壓Vth_max時，比較器CMP3輸出高電平，因此或閘G2輸出的控制信號s2為高電平，在此情況下，開關Q關斷。 According to the circuit shown in FIG. 7, when the voltage Vcs on the current detecting resistor reaches the first overcurrent protection threshold voltage Vth_max, the comparator CMP3 outputs a high level, so that the control signal s2 outputted by the gate G2 is at a high level. In this case, the switch Q is turned off.

當電池的充電電壓Vo小於電壓V1時，採樣電壓Vc0小於參考電壓Vref1，此時誤差放大器EA1輸出的電壓信號Vea1為電源電壓AVDD。在本發明中，設定斜坡信號Vramp的電壓範圍為V3~V4(其中V3<V4<AVDD)，同時設定第一過電流保護閾值電壓Vth_max為： ，因此當誤差放大器EA1輸出的電壓信號Vea1的電壓值高於V4時，電壓控制信號Vctrl的電壓(也即)高於第一過電流保護閾值電壓Vth_max。因此，電流檢測電阻Rs上的電壓信號Vcs小於第一過電流保護閾值電壓Vth_max時，比較器CMP1對電壓控制信號Vctrl和電流檢測電阻Rs上的電壓信號Vcs進行比較的結果為低電平，因此及閘A4輸出的信號為低電平，此時比較器CMP3輸出的比較結果也為低電平，因此或閘G2輸出的控制信號s2為低電平；在電流檢測電阻Rs上的電壓信號Vcs等於第一過電流保護閾值電壓Vth_max時，比較器CMP3輸出高電平，因此或閘G2輸出的控制信號s2為高電平。 When the charging voltage Vo of the battery is less than the voltage V1, the sampling voltage Vc0 is smaller than the reference voltage Vref1, and the voltage signal Vea1 output by the error amplifier EA1 is the power supply voltage AVDD. In the present invention, the voltage range of the ramp signal Vramp is set to V3 to V4 (where V3 < V4 < AVDD), and the first overcurrent protection threshold voltage Vth_max is set to: Therefore, when the voltage value of the voltage signal Vea1 outputted by the error amplifier EA1 is higher than V4, the voltage of the voltage control signal Vctrl (ie, ) is higher than the first overcurrent protection threshold voltage Vth_max. Therefore, when the voltage signal Vcs on the current detecting resistor Rs is smaller than the first overcurrent protection threshold voltage Vth_max, the comparator CMP1 compares the voltage control signal Vctrl with the voltage signal Vcs on the current detecting resistor Rs to a low level. The signal output from the gate A4 is low level. At this time, the comparison result of the output of the comparator CMP3 is also low level, so the control signal s2 outputted by the gate G2 is low level; the voltage signal Vcs on the current detecting resistor Rs When the first overcurrent protection threshold voltage Vth_max is equal, the comparator CMP3 outputs a high level, and thus the control signal s2 outputted by the gate G2 is at a high level.

當電池的充電電壓Vo等於電壓V1時，採樣電壓Vc0等於參考電壓Vref1，根據本發明示例性實施例的反激式電源變換系統工作在第一恒壓充電控制模式下，誤差放大器EA1輸出的電壓信號Vea1在電壓V3~V4之間 變化，因此電壓控制信號Vctrl的電壓也在和之間變化。當電流檢測電阻Rs上的電壓信號Vcs等於電壓控制信號Vctrl和第二過電流保護閾值電壓Vth_min中電壓比較高的電壓時，比較器CMP1和CMP2同時輸出高電平，及閘A4輸出為1，從而或閘G2輸出的控制信號s2為高電平。另外，當電流檢測電阻Rs上的電壓信號Vcs到達第一過電流保護閾值電壓Vth_max時，比較器CMP3也輸出高電平信號，因而或閘G2輸出的控制信號s2也為高電平。也就是說，在第一恒壓控制模式下，電壓控制信號Vctrl和第二過電流保護閾值電壓Vth_min中比較高的電壓和第一過電流保護閾值電壓Vth_max比較，選擇一個比較低的電壓，當電流檢測電阻Rs上的電壓信號Vcs等於該較低的電壓時，或閘G2輸出高電平，開關Q關斷。另外在電流檢測電阻Rs上的電壓信號Vcs小於電壓控制信號Vctrl和第二過電流保護閾值電壓Vth_min中的至少一個並且小於第一過電流保護閾值電壓Vth_max時，或閘G2輸出的控制信號s2為低電平。 When the charging voltage Vo of the battery is equal to the voltage V1, the sampling voltage Vc0 is equal to the reference voltage Vref1, and the flyback power conversion system according to an exemplary embodiment of the present invention operates in the first constant voltage charging control mode, and the voltage output by the error amplifier EA1 The signal Vea1 varies between voltages V3 and V4, so the voltage of the voltage control signal Vctrl is also with Change between. When the voltage signal Vcs on the current detecting resistor Rs is equal to the voltage of the voltage control signal Vctrl and the second overcurrent protection threshold voltage Vth_min, the comparators CMP1 and CMP2 simultaneously output a high level, and the gate A4 output is 1, Thus, the control signal s2 output from the gate G2 is at a high level. Further, when the voltage signal Vcs on the current detecting resistor Rs reaches the first overcurrent protection threshold voltage Vth_max, the comparator CMP3 also outputs a high level signal, and thus the control signal s2 output from the gate G2 is also at a high level. That is, in the first constant voltage control mode, the relatively high voltage of the voltage control signal Vctrl and the second overcurrent protection threshold voltage Vth_min is compared with the first overcurrent protection threshold voltage Vth_max, and a relatively low voltage is selected. When the voltage signal Vcs on the current detecting resistor Rs is equal to the lower voltage, or the gate G2 outputs a high level, the switch Q is turned off. Further, when the voltage signal Vcs on the current detecting resistor Rs is smaller than at least one of the voltage control signal Vctrl and the second overcurrent protection threshold voltage Vth_min and smaller than the first overcurrent protection threshold voltage Vth_max, the control signal s2 output by the gate G2 is Low level.

當電池的充電電壓Vo大於電壓V1而小於電壓V2時(即當採樣電壓Vc0大於參考電壓Vref1而小於參考電壓Vref2時)，根據本發明示例性實施例的反激式電源變換系統工作在第二恒流充電控制模式，而在電池的充電電壓Vo等於電壓V2時(即當採樣電壓Vco等於參考電壓Vref2時)，根據本發明示例性實施例的反激式電源變換系統工作在第二恒壓充電控制模式下。在這兩種情況下，誤差放大器EA1開環並輸出低電平信號，因此比較器CMP1輸出高電平信號，當電流檢測電阻Rs上的電壓信號Vcs的電壓值小於第二過電流保護閾值電壓Vth_min(因此也小於第一過電流保護閾值電壓Vth_max)時，或閘G2輸出為低電平信號；而當電流檢測電阻Rs上的電壓信號Vcs的電壓值等於第二過電流保護閾值電壓Vth_min時，或閘G2輸出高電平信號，開關Q關斷。 When the charging voltage Vo of the battery is greater than the voltage V1 and less than the voltage V2 (ie, when the sampling voltage Vc0 is greater than the reference voltage Vref1 and smaller than the reference voltage Vref2), the flyback power conversion system according to an exemplary embodiment of the present invention operates in the second The constant current charging control mode, and when the charging voltage Vo of the battery is equal to the voltage V2 (ie, when the sampling voltage Vco is equal to the reference voltage Vref2), the flyback power conversion system according to an exemplary embodiment of the present invention operates at the second constant voltage In charge control mode. In both cases, the error amplifier EA1 opens and outputs a low level signal, so the comparator CMP1 outputs a high level signal when the voltage value of the voltage signal Vcs on the current detecting resistor Rs is less than the second overcurrent protection threshold voltage. When Vth_min (and therefore also less than the first overcurrent protection threshold voltage Vth_max), OR gate G2 outputs a low level signal; and when the voltage value of the voltage signal Vcs on the current detecting resistor Rs is equal to the second overcurrent protection threshold voltage Vth_min , or gate G2 outputs a high level signal, switch Q is turned off.

第8圖示出了根據本發明示例性實施例的充電控制電路530 的邏輯驅動電路5320的電路示意圖。應當理解，第8圖所示的電路只是為了便於本領域技術人員更好地理解本發明，而並不作為本發明的一種限制。 FIG. 8 illustrates a charge control circuit 530 in accordance with an exemplary embodiment of the present invention. A schematic diagram of the logic drive circuit 5320. It should be understood that the circuit shown in FIG. 8 is only for the purpose of facilitating a better understanding of the present invention by those skilled in the art, and is not a limitation of the present invention.

根據本發明的示例性實施例，所述邏輯驅動電路5320包括兩個反或閘NG1和NG2的輸入和輸出交叉耦合而成的觸發電路和與所述觸發電路相連的驅動器。其中，反或閘NG1的一個輸入為從模式選擇電路5310輸出的控制信號s1，其另一輸入為反或閘NG2輸出的電平信號；反或閘NG2的一個輸入為從過電流保護電路5330輸出的控制信號s2，其另一輸入為反或閘NG1輸出的電平信號。觸發電路中的反或閘NG2輸出的電平信號輸入至驅動器GD8，驅動器GD8輸出驅動控制信號Sdrv以驅動開關Q。這裡，當控制信號s1為高電平而控制信號s2為低電平時，觸發器輸出為高電平，從而使驅動器輸出驅動控制信號Sdrv為高電平信號進而以使開關Q導通。 According to an exemplary embodiment of the present invention, the logic driving circuit 5320 includes a flip-flop circuit in which two inputs or outputs of the inverse gates NG1 and NG2 are cross-coupled and a driver connected to the flip-flop circuit. Wherein, one input of the inverse gate NG1 is the control signal s1 outputted from the mode selection circuit 5310, and the other input is the level signal output by the inverse gate NG2; one input of the inverse gate NG2 is the slave overcurrent protection circuit 5330 The output control signal s2, the other input of which is the level signal of the inverse or gate NG1 output. A level signal of the inverse or gate NG2 output in the flip-flop circuit is input to the driver GD8, and the driver GD8 outputs a drive control signal Sdrv to drive the switch Q. Here, when the control signal s1 is at a high level and the control signal s2 is at a low level, the flip-flop output is at a high level, thereby causing the driver to output the drive control signal Sdrv to a high level signal to turn on the switch Q.

本領域技術人員應該理解，驅動開關Q的驅動電路並不限於第8圖所示的電路結構，其可以是任何可以實現所述功能的驅動電路，而本發明所示出的電路結構僅用作示例的目的。 It should be understood by those skilled in the art that the driving circuit for driving the switch Q is not limited to the circuit structure shown in FIG. 8, which may be any driving circuit that can realize the function, and the circuit structure shown in the present invention is only used as the circuit structure. The purpose of the example.

下面將參照第9圖至第12圖來更加詳細地描述上述四種充電控制模式。這裡需要注意的是，根據本發明示例性實施例的第一恒壓充電控制模式表示反激式電源變換系統在電壓控制信號Vctrl的控制下，電流檢測電阻Rs上的峰值電壓Vcs_peak在第二過電流保護閾值電壓Vth_min和第一過電流保護閾值電壓Vth_max之間變化，這時通過脈波頻率調變模式(PFM)實現電池的充電電壓的恒定；根據本發明示例性實施例的第二恒壓充電控制模式也僅示意此時反激式電源變換系統由第二過電流保護閾值電壓Vth_min和脈衝頻率模式(PFM)下實現恒壓功能。本領域技術人員應該理解，下面給出的電路結構只是本發明的一個示例，本發明並不限於如下描述的具體的電路結構，在不脫離本發明的範圍的情況下本領域技術人員可以採用其他電路結構來實現。 The above four charging control modes will be described in more detail below with reference to FIGS. 9 to 12. It should be noted here that the first constant voltage charging control mode according to an exemplary embodiment of the present invention indicates that the flyback power conversion system is under the control of the voltage control signal Vctrl, and the peak voltage Vcs_peak on the current detecting resistor Rs is in the second The current protection threshold voltage Vth_min and the first overcurrent protection threshold voltage Vth_max are varied, at which time the charging voltage of the battery is constant by the pulse wave frequency modulation mode (PFM); the second constant voltage charging according to an exemplary embodiment of the present invention The control mode also only indicates that the flyback power conversion system achieves a constant voltage function by the second overcurrent protection threshold voltage Vth_min and the pulse frequency mode (PFM). It should be understood by those skilled in the art that the circuit structure given below is only an example of the present invention, and the present invention is not limited to the specific circuit structure described below, and those skilled in the art may adopt other embodiments without departing from the scope of the present invention. The circuit structure is implemented.

第一恒流充電控制模式 First constant current charging control mode

在本說明書中，電平信號CC_high、CC_low、CV1以及CV2都是由高電平“1”和低電平“0”構成的信號。當電池的充電電壓Vo小於電壓V1時，由於採樣電壓Vc0小於參考電壓Vref1，因此誤差放大器EA1輸出的電壓信號Vea1為電源電壓AVDD，這裡設定斜坡信號發生器輸出的斜坡信號Vramp電壓的範圍為V3~V4，其中V3<V4<AVDD，因此比較器CMP0輸出一直為高電平。另外，在電池的充電電壓Vo小於電壓V2時，即在採樣電壓Vc0小於參考電壓Vref2時，電平信號CV2輸出為高電平，而電平信號CC_low的頻率低於電平信號CC_high的頻率，因此或閘G1輸出的控制信號s1的頻率由電平信號CC_high決定。反激式電源變換系統工作在第一恒流充電控制模式。 In the present specification, the level signals CC_high, CC_low, CV1, and CV2 are signals composed of a high level "1" and a low level "0". When the charging voltage Vo of the battery is less than the voltage V1, since the sampling voltage Vc0 is smaller than the reference voltage Vref1, the voltage signal Vea1 output by the error amplifier EA1 is the power supply voltage AVDD, and the range of the ramp signal Vramp voltage output by the ramp signal generator is set to V3. ~V4, where V3<V4<AVDD, so the comparator CMP0 output is always high. In addition, when the charging voltage Vo of the battery is less than the voltage V2, that is, when the sampling voltage Vc0 is smaller than the reference voltage Vref2, the level signal CV2 is outputted to a high level, and the frequency of the level signal CC_low is lower than the frequency of the level signal CC_high, Therefore, the frequency of the control signal s1 output by the gate G1 is determined by the level signal CC_high. The flyback power conversion system operates in a first constant current charging control mode.

第9圖是根據本發明示例性實施例的第一恒流充電控制電路640的示例性電路圖。 FIG. 9 is an exemplary circuit diagram of a first constant current charging control circuit 640 according to an exemplary embodiment of the present invention.

如第9圖所示，所述第一恒流充電控制電路640由電流鏡I1(為了方便閱讀，其輸出電流也用I1表示)和電流鏡I1’、反閘N9、電容C9、比較器cmp_cc以及兩個開關K1和K1’構成，其中，電流鏡I1’的電流是電流鏡I1的輸出電流的k1倍。這裡，開關K1’和電流鏡I1’串聯形成串聯電路，該串聯電路與電容C9並聯形成並聯電路。電流鏡I1、開關K1依次與上述並聯電路一端相連，而該並聯電路的另一端接地。其中，電流鏡I1的另一端接到電源電壓AVDD。這裡從退磁檢測器620輸出的退磁信號demag控制開關K1’的斷開和閉合，同時該退磁信號demag經過反閘N9來控制開關K1的斷開和閉合。電容C9上的電壓信號Vramp_cc輸入到比較器cmp_cc的正向輸入端，而參考電壓Va則輸入到比較器cmp_cc的負向輸入端以進行比較。當電壓信號Vramp_cc的電壓值大於或等於參考電壓Va時，比較器cmp_cc輸出的電平信號CC_high為高電平，否則為低電平。 As shown in FIG. 9, the first constant current charging control circuit 640 is composed of a current mirror I1 (the output current is also represented by I1 for convenience of reading) and a current mirror I1', a reverse gate N9, a capacitor C9, and a comparator cmp_cc. And two switches K1 and K1', wherein the current of the current mirror I1' is k1 times the output current of the current mirror I1. Here, the switch K1' and the current mirror I1' are connected in series to form a series circuit which is connected in parallel with the capacitor C9 to form a parallel circuit. The current mirror I1 and the switch K1 are sequentially connected to one end of the parallel circuit, and the other end of the parallel circuit is grounded. The other end of the current mirror I1 is connected to the power supply voltage AVDD. Here, the demagnetization signal demag output from the demagnetization detector 620 controls the opening and closing of the switch K1', while the demagnetization signal demag passes through the reverse gate N9 to control the opening and closing of the switch K1. The voltage signal Vramp_cc on the capacitor C9 is input to the forward input terminal of the comparator cmp_cc, and the reference voltage Va is input to the negative input terminal of the comparator cmp_cc for comparison. When the voltage value of the voltage signal Vramp_cc is greater than or equal to the reference voltage Va, the level signal CC_high output by the comparator cmp_cc is a high level, and otherwise is a low level.

根據第9圖所示的電路，當退磁信號demag為高電平(即系統工作在退磁狀態)時，開關K1’閉合，開關K1斷開，電流鏡I1’對電容C9放電，電壓信號Vramp_cc的電壓線性下降。當退磁信號demag為低電平時， 開關K1閉合，開關K1’斷開，電流鏡I1對電容C9充電；此時，當電壓信號Vramp_cc的電壓高於參考電壓Va時，比較器cmp_cc輸出的電平信號CC_high為高電平，從而可控制開關Q的導通。 According to the circuit shown in Fig. 9, when the demagnetization signal demag is at a high level (that is, the system operates in a demagnetization state), the switch K1' is closed, the switch K1 is turned off, the current mirror I1' is discharged to the capacitor C9, and the voltage signal Vramp_cc is The voltage drops linearly. When the demagnetization signal demag is low, The switch K1 is closed, the switch K1' is turned off, and the current mirror I1 charges the capacitor C9; at this time, when the voltage of the voltage signal Vramp_cc is higher than the reference voltage Va, the level signal CC_high outputted by the comparator cmp_cc is high level, thereby Controls the conduction of the switch Q.

第10圖是根據本發明示例性實施例的第一恒流充電控制模式下的時序圖，其中，橫軸表示時間。V_FB表示在輔助繞組Naux上的FB處的回饋電壓，Vd表示退磁檢測器620檢測退磁過程時使用的參考電壓。當回饋電壓V_FB大於該參考電壓時，退磁檢測器620輸出高電平的退磁信號demag，而當回饋電壓V_FB低於該參考電壓時，退磁檢測器620判斷退磁結束而輸出低電平的退磁信號demag。Tdemag表示退磁信號demag為高電平的時間區間。關於第9圖中的電容C9上的電壓信號Vramp_cc，可以看到在退磁過程期間該電壓信號Vramp_cc的電壓逐漸降低，而在退磁信號demag為低電平時，其電壓逐漸升高。其中，在電壓信號Vramp_cc的電壓高於參考電壓Va時，電平信號High_cc為高電平。另外第10圖中的Sdrv是用以表示開關Q的導通和關斷的驅動控制信號。當驅動控制信號Sdrv為高電平時，開關Q導通，而當驅動控制信號Sdrv為低電平時，開關Q關斷。這裡在開關Q導通時，電流檢測電阻Rs上的電壓信號Vcs逐漸上升，而當電壓信號Vcs上升到第一過電流保護閾值電壓Vth_max時，驅動控制信號Sdrv變為低電平，開關Q關斷。這裡，第10圖中的Tcc_h表示開關Q在第一恒流充電控制模式時的操作週期。 Fig. 10 is a timing chart in the first constant current charging control mode according to an exemplary embodiment of the present invention, wherein the horizontal axis represents time. V _FB represents the feedback voltage at the _FB on the auxiliary winding Naux, and Vd represents the reference voltage used when the demagnetization detector 620 detects the demagnetization process. When the feedback voltage V _{FB is} greater than the reference voltage, the demagnetization detector 620 outputs a high level demagnetization signal demag, and when the feedback voltage V _{FB is} lower than the reference voltage, the demagnetization detector 620 determines that the demagnetization is ended and outputs a low level. Demagnetization signal demag. Tdemag represents the time interval during which the demagnetization signal demag is high. Regarding the voltage signal Vramp_cc on the capacitor C9 in Fig. 9, it can be seen that the voltage of the voltage signal Vramp_cc gradually decreases during the demagnetization process, and the voltage gradually rises when the demagnetization signal demag is low. Wherein, when the voltage of the voltage signal Vramp_cc is higher than the reference voltage Va, the level signal High_cc is at a high level. Further, Sdrv in Fig. 10 is a drive control signal for indicating ON and OFF of the switch Q. When the drive control signal Sdrv is at a high level, the switch Q is turned on, and when the drive control signal Sdrv is at a low level, the switch Q is turned off. Here, when the switch Q is turned on, the voltage signal Vcs on the current detecting resistor Rs gradually rises, and when the voltage signal Vcs rises to the first overcurrent protection threshold voltage Vth_max, the driving control signal Sdrv becomes a low level, and the switch Q is turned off. . Here, Tcc_h in Fig. 10 indicates an operation period of the switch Q in the first constant current charge control mode.

由上面分析可以知道，第9圖中的電容C9上電壓差為： From the above analysis, we can know that the voltage difference on capacitor C9 in Figure 9 is:

因此，開關的工作週期Tcc_h為： Therefore, the duty cycle Tcc_h of the switch is:

變壓器原邊側的峰值電流為： The peak current on the primary side of the transformer is:

假設變壓器傳輸效率為100%，則在第一恒流充電控制模式下對電池進行充電的電流，也即副邊的輸出電流為： Assuming that the transformer transmission efficiency is 100%, the current that charges the battery in the first constant current charging control mode, that is, the output current of the secondary side is:

其中，N為變壓器原邊匝數和副邊匝數的比值，k1為固定值，Vth_max為第一過電流保護閥值電壓，Rs為電流檢測電阻的阻值。 Where N is the ratio of the primary and secondary turns of the transformer, k1 is a fixed value, Vth_max is the first overcurrent protection threshold voltage, and Rs is the resistance of the current sense resistor.

第一恒壓充電控制模式First constant voltage charging control mode

當電池的充電電壓Vo等於電壓V1時，由於採樣電壓Vc0等於參考電壓Vref1，因此誤差放大器EA1輸出的電壓信號Vea1的電壓在V3~V4之間變化，如上所述，V3<V4<AVDD。此時，反激式電源變換系統工作在第一恒壓充電控制模式，誤差放大器EA1輸出的電壓信號Vea1的大小反應了輸出電流大小，反激式電源變換系統的工作頻率(即開關Q的工作頻率)，由電平信號CV1的頻率Fcv1決定。具體來講，從誤差放大器EA1輸出的電壓信號Vea1輸入到比較器CMP0的正向輸入端，而從斜坡信號發生器630輸出的斜坡信號Vramp輸入到比較器CMP0的反向輸入端，誤差放大器EA1輸出的電壓信號Vea1的電壓越低，則在每個工作週期內，斜坡信號Vramp的電壓值大於電壓信號Vea1的電壓的時間越長，因此引起比較器CMP0輸出的電平信號CV1的頻率Fcv1越低。其中，斜坡信號發生器630輸出的斜坡信號Vramp的電壓範圍在V3-V4之間變化。為了便於本領域技術人員更好地理解這一點兒，在此將對其進行示例性的描述。當退磁信號demag變為高電平時，斜坡信號Vramp就被重定到初始值電壓V4。舉例來說，假設斜坡信號Vramp的電壓呈線性變化，V3=3V，V4=1V，從電壓V3下降到電壓V4的時間為1mS，假設此時誤差放大器EA1輸出的電壓信號Vea1的電壓為2V，則斜坡信號 Vramp的電壓下降到2V時，比較器CMP0輸出為“1”，開關Q可以導通，而當開關Q關斷時，退磁信號demag就又變成高電平。在這種情況下，斜坡信號Vramp從重定值3V下降到2V的時間只有0.5ms。因此，比較器CMP0輸出低電平的時間等於斜坡信號從初始值電壓V4下降到誤差放大器EA1輸出的電壓信號Vea1時間，高電平時間等於DRV開啟時間。因此說在反激式電源變換系統工作在第一恒壓充電控制模式時，電壓信號Vea1的電壓越低，則斜坡信號Vramp下降到該電壓信號Vea1的電壓所用時間越長，電平信號CV1輸出頻率越低。 When the charging voltage Vo of the battery is equal to the voltage V1, since the sampling voltage Vc0 is equal to the reference voltage Vref1, the voltage of the voltage signal Vea1 output by the error amplifier EA1 varies between V3 and V4, as described above, V3 < V4 < AVDD. At this time, the flyback power conversion system operates in the first constant voltage charging control mode, and the magnitude of the voltage signal Vea1 outputted by the error amplifier EA1 reflects the magnitude of the output current, and the operating frequency of the flyback power conversion system (ie, the operation of the switch Q) The frequency is determined by the frequency Fcv1 of the level signal CV1. Specifically, the voltage signal Vea1 output from the error amplifier EA1 is input to the forward input terminal of the comparator CMP0, and the ramp signal Vramp output from the ramp signal generator 630 is input to the inverting input terminal of the comparator CMP0, the error amplifier EA1 The lower the voltage of the output voltage signal Vea1, the longer the voltage value of the ramp signal Vramp is greater than the voltage of the voltage signal Vea1 in each duty cycle, so the frequency Fcv1 of the level signal CV1 output by the comparator CMP0 is increased. low. The voltage range of the ramp signal Vramp output by the ramp signal generator 630 varies between V3-V4. In order to facilitate a person skilled in the art to better understand this, it will be exemplarily described herein. When the demagnetization signal demag becomes a high level, the ramp signal Vramp is reset to the initial value voltage V4. For example, suppose the voltage of the ramp signal Vramp changes linearly, V3=3V, V4=1V, and the time from voltage V3 to voltage V4 is 1mS, assuming that the voltage of the voltage signal Vea1 output by the error amplifier EA1 is 2V. Ramp signal When the voltage of Vramp drops to 2V, the output of the comparator CMP0 is "1", the switch Q can be turned on, and when the switch Q is turned off, the demagnetization signal demag becomes high again. In this case, the ramp signal Vramp drops from the reset value of 3V to 2V for only 0.5 ms. Therefore, the time during which the comparator CMP0 outputs a low level is equal to the time when the ramp signal falls from the initial value voltage V4 to the voltage signal Vea1 output by the error amplifier EA1, and the high level time is equal to the DRV turn-on time. Therefore, when the flyback power conversion system operates in the first constant voltage charging control mode, the lower the voltage of the voltage signal Vea1, the longer the ramp signal Vramp falls to the voltage of the voltage signal Vea1, and the level signal CV1 is output. The lower the frequency.

電平信號CV1和CC_high經過及閘A2進行邏輯與操作，因此電平信號CV1和CC_high中頻率低的一個信號被選擇為及閘A2的輸出信號。當電壓信號Vea1低到一定程度時，電平信號CV1的頻率Fcv1小於電平信號CC_high的頻率Fcc_high，因此及閘A2輸出的電平信號的頻率就由電平信號CV1的頻率Fcv1決定，此時電平信號CV1的頻率Fcv1低於電平信號CC_high的頻率Fcc_high而高於電平信號CC_low的頻率Fcc_low。因此，或閘G1輸出的控制信號s1由電平信號CV1來決定。此時，反激式電源變換系統工作在第一恒壓充電控制模式。 The level signals CV1 and CC_high are logically ANDed by the AND gate A2, so that a signal having a low frequency among the level signals CV1 and CC_high is selected as the output signal of the AND gate A2. When the voltage signal Vea1 is low to a certain extent, the frequency Fcv1 of the level signal CV1 is smaller than the frequency Fcc_high of the level signal CC_high, and therefore the frequency of the level signal outputted by the gate A2 is determined by the frequency Fcv1 of the level signal CV1. The frequency Fcv1 of the level signal CV1 is lower than the frequency Fcc_high of the level signal CC_high and higher than the frequency Fcc_low of the level signal CC_low. Therefore, the control signal s1 output from the gate G1 is determined by the level signal CV1. At this time, the flyback power conversion system operates in the first constant voltage charging control mode.

第二恒流充電控制模式Second constant current charging control mode

當電池的充電電壓Vo大於電壓V1而小於電壓V2時，採樣電壓Vc0的電壓大於參考電壓Vref1而小於參考電壓Vref2，因此誤差放大器EA1輸出的電壓信號Vea1的電壓為0。因此，從比較器CMP0輸出的電平信號CV1為低電平，及閘A2輸出的電壓信號為低電平，也即或閘G1的輸出信號由及閘A3輸出的信號決定。因為此時與第二恒流充電控制電路650輸出的電平信號CC_low進行邏輯與操作的電平信號CV2依然為高電平，因此或閘G1輸出的控制信號s1由電平信號CC_low決定。此時，反激式電源變換系統工作在第二恒流充電控制模式。 When the charging voltage Vo of the battery is greater than the voltage V1 and smaller than the voltage V2, the voltage of the sampling voltage Vc0 is greater than the reference voltage Vref1 and smaller than the reference voltage Vref2, so the voltage of the voltage signal Vea1 output by the error amplifier EA1 is zero. Therefore, the level signal CV1 outputted from the comparator CMP0 is at a low level, and the voltage signal output from the gate A2 is at a low level, that is, the output signal of the gate G1 is determined by the signal output from the gate A3. Since the level signal CV2 that is logically AND-operated with the level signal CC_low outputted by the second constant current charging control circuit 650 is still at the high level at this time, the control signal s1 output from the gate G1 is determined by the level signal CC_low. At this time, the flyback power conversion system operates in the second constant current charging control mode.

第11圖是根據本發明示例性實施例的第二恒流充電控制電 路650的示例性電路圖。本領域技術人員應該理解，該電路圖僅是為了使本領域技術人員更加容易地理解本發明而給出的示例，並不作為對本發明的限制。 11 is a second constant current charging control power according to an exemplary embodiment of the present invention An exemplary circuit diagram of way 650. It will be understood by those skilled in the art that the present invention is only an example of the present invention and is not intended to limit the invention.

如第11圖所示，所述第二恒流充電控制電路650由電流鏡I2和電流鏡I2’、反閘N11、電容C11、比較器cmp_ccl以及兩個開關K2和K2’構成，其中，電流鏡I2’的電流是電流鏡I2的電流的k2(k2>k1)倍。這裡，開關K2’和電流鏡I2’串聯形成串聯電路，該串聯電路與電容C11並聯形成並聯電路。電流鏡I2、開關K2串聯之後與上述並聯電路一端相連，而該並聯電路的另一端接地。其中，電流鏡I2的另一端輸入電源電壓AVDD。電容C11上的電壓信號Vramp_ccl輸入到比較器cmp_ccl的正向輸入端，而參考電壓Va則輸入到比較器cmp_ccl的負向輸入端。當電壓信號Vramp_ccl的電壓值大於或等於參考電壓Va時，比較器cmp_ccl輸出的電平信號CC_low為高電平，否則為低電平。這裡從退磁檢測器620輸出的退磁信號demag直接控制開關K2’的斷開和閉合，同時經過反閘N11來控制開關K2的斷開和閉合。 As shown in FIG. 11, the second constant current charging control circuit 650 is composed of a current mirror I2 and a current mirror I2', a reverse gate N11, a capacitor C11, a comparator cmp_ccl, and two switches K2 and K2', wherein the current The current of the mirror I2' is k2 (k2>k1) times the current of the current mirror I2. Here, the switch K2' and the current mirror I2' are connected in series to form a series circuit which is connected in parallel with the capacitor C11 to form a parallel circuit. The current mirror I2 and the switch K2 are connected in series to one end of the parallel circuit, and the other end of the parallel circuit is grounded. The other end of the current mirror I2 is input with a power supply voltage AVDD. The voltage signal Vramp_ccl on the capacitor C11 is input to the forward input terminal of the comparator cmp_ccl, and the reference voltage Va is input to the negative input terminal of the comparator cmp_ccl. When the voltage value of the voltage signal Vramp_ccl is greater than or equal to the reference voltage Va, the level signal CC_low output by the comparator cmp_ccl is a high level, and otherwise is a low level. Here, the demagnetization signal demag outputted from the demagnetization detector 620 directly controls the opening and closing of the switch K2' while controlling the opening and closing of the switch K2 via the reverse gate N11.

可以看到第11圖所示的電路與第9圖所示的電路具有基本上相同的電路結構，因此在這裡不對第11圖所示的電路的工作原理進行詳細描述。這裡需要注意的是，在第二恒流充電控制模式下，變壓器原邊峰值電流為： It can be seen that the circuit shown in Fig. 11 has substantially the same circuit structure as the circuit shown in Fig. 9, and therefore the operation principle of the circuit shown in Fig. 11 is not described in detail herein. It should be noted here that in the second constant current charging control mode, the primary peak current of the transformer is:

基於上面對第一恒流充電控制模式下電池的充電電流，也即副邊繞組的輸出電流的推導過程及分析，可知在第二恒流充電控制模式下副邊繞組的輸出電流Icc_l為： Based on the above-mentioned derivation process and analysis of the charging current of the battery in the first constant current charging control mode, that is, the output current of the secondary winding, it can be seen that the output current Icc_l of the secondary winding in the second constant current charging control mode is:

因此，由公式(11)和公式(13)可以得到： 其中，k2>k1>1，並且k1和k2都為定值。 Therefore, from equation (11) and formula (13), we can get: Where k2>k1>1 and both k1 and k2 are fixed values.

通常，第一過電流保護閾值電壓Vth_max是給定值，第二過電流保護閾值電壓Vth_min的正常取值範圍一般在1/3~1/2倍的第一過電流保護閾值電壓Vth_max。當然上面給出的範圍只是示例性的，根據本發明的實施例，通常設定k1為固定值(例如k1=1.5)，根據電池特性所需要的Icc_h和Icc_l比值要求，最後計算出k2；也可以預先設定好k1、k2，計算出第二過電流保護閾值電壓Vth_min的取值。這裡，為了避免採樣出錯，第二過電流保護閾值電壓Vth_min通常不能小於1/3倍的第一過電流保護閾值電壓Vth_max。 Generally, the first overcurrent protection threshold voltage Vth_max is a given value, and the normal value range of the second overcurrent protection threshold voltage Vth_min is generally 1/3 to 1/2 times the first overcurrent protection threshold voltage Vth_max. Of course, the ranges given above are merely exemplary. According to an embodiment of the present invention, k1 is usually set to a fixed value (for example, k1=1.5), and the ratio of Icc_h and Icc_l required according to battery characteristics is finally calculated, and finally k2 is calculated; The k1 and k2 are set in advance, and the value of the second overcurrent protection threshold voltage Vth_min is calculated. Here, in order to avoid sampling errors, the second overcurrent protection threshold voltage Vth_min is generally not less than 1/3 times the first overcurrent protection threshold voltage Vth_max.

第二恒壓充電控制模式Second constant voltage charging control mode

當電池的充電電壓Vo等於電壓V2時(即採樣電壓Vc0等於參考電壓Vref2時)，採樣電壓Vc0大於參考電壓Vref1，因此誤差放大器EA1輸出的電壓信號Vea1的電壓為0。因此，從比較器CMP0輸出的電平信號CV1為低電平，及閘A2輸出的電壓信號為低電平，也即或閘G1的輸出信號由及閘A3輸出的信號決定。因為此時與第二恒流充電控制電路650輸出的電平信號CC_low進行邏輯與操作的電平信號CV2的頻率Fcv2低於電平信號CC_low的頻率Fcc_low，因此或閘G1輸出的控制信號s1由電平信號CV2決定。此時，反激式電源變換系統工作在第二恒壓充電控制狀態。 When the charging voltage Vo of the battery is equal to the voltage V2 (ie, when the sampling voltage Vc0 is equal to the reference voltage Vref2), the sampling voltage Vc0 is greater than the reference voltage Vref1, and thus the voltage of the voltage signal Vea1 output by the error amplifier EA1 is zero. Therefore, the level signal CV1 outputted from the comparator CMP0 is at a low level, and the voltage signal output from the gate A2 is at a low level, that is, the output signal of the gate G1 is determined by the signal output from the gate A3. Since the frequency Fcv2 of the level signal CV2 that is logically ANDed with the level signal CC_low outputted by the second constant current charging control circuit 650 is lower than the frequency Fcc_low of the level signal CC_low, the control signal s1 output by the gate G1 is The level signal CV2 is determined. At this time, the flyback power conversion system operates in the second constant voltage charging control state.

第12圖示出了根據本發明示例性實施例的模式選擇電路5310中的第二恒壓充電控制電路660示例性電路圖。如第12圖所示，所述第二恒壓充電控制電路660包括誤差放大器EA2、電容C20、壓控振盪器VCO。其中，誤差放大器EA2的負向輸入端輸入採樣電壓Vc0，其正向輸入端輸入 參考電壓Vref2，其中，當電池的充電電壓到達電壓V2時，採樣電壓Vc0等於參考電壓Vref2，這裡，參考電壓Vref2大於參考電壓Vref1。誤差放大器EA2的輸出端分別連接到壓控振盪器VCO和電容C20，其中，電容C20的另一端接地。誤差放大器EA2對輸入的信號之間的電壓差進行放大並輸出電壓信號Vea2並向壓控振盪器VCO輸入電壓信號Vea2。壓控振盪器VCO在電壓信號Vea2的控制下輸出具有某種頻率的電平信號CV2。 FIG. 12 shows an exemplary circuit diagram of a second constant voltage charge control circuit 660 in the mode selection circuit 5310 according to an exemplary embodiment of the present invention. As shown in FIG. 12, the second constant voltage charging control circuit 660 includes an error amplifier EA2, a capacitor C20, and a voltage controlled oscillator VCO. Wherein, the negative input terminal of the error amplifier EA2 inputs the sampling voltage Vc0, and its positive input terminal inputs The reference voltage Vref2, wherein when the charging voltage of the battery reaches the voltage V2, the sampling voltage Vc0 is equal to the reference voltage Vref2, where the reference voltage Vref2 is greater than the reference voltage Vref1. The output of the error amplifier EA2 is connected to the voltage controlled oscillator VCO and the capacitor C20, respectively, wherein the other end of the capacitor C20 is grounded. The error amplifier EA2 amplifies the voltage difference between the input signals and outputs the voltage signal Vea2 and inputs the voltage signal Vea2 to the voltage controlled oscillator VCO. The voltage controlled oscillator VCO outputs a level signal CV2 having a certain frequency under the control of the voltage signal Vea2.

第13圖示出了壓控振盪器VCO的輸出頻率Fcv2與電壓信號Vea2之間的關係示圖。如第13圖所示，在電壓信號Vea2小於電壓Vg時，壓控振盪器VCO輸出的電平信號CV2的頻率Fcv2為頻率Fmin，在電壓信號Vea2大於電壓Vf而小於電壓Ve時，壓控振盪器VCO輸出的電平信號CV2的頻率Fcv2為Fcc_low，而電壓信號Vea2在電壓Vg和電壓Vf之間變化時，壓控振盪器VCO輸出的電平信號CV2的頻率Fcv2在從頻率Fmin逐漸上升到頻率Fcc_low，當電壓信號Vea2的電壓大於電壓Ve時，電平信號CV2一直為高電平，也即電平信號CV2的頻率Fcv2為0。因此，電平信號CV2的頻率Fcv2也隨著電池的充電電壓Vo而增大，從頻率為0變為頻率Fcc_low，然後又下降至頻率Fmin。當電池的充電電壓Vo等於電壓V2時，電壓信號Vea2控制壓控振盪器VCO輸出頻率Fcv2小於頻率Fcc_low和頻率Fcc_high，因此頻率Fcc_low和頻率Fcc_high被遮罩，控制信號s1由電平信號CV2決定，電流檢測電阻Rs上的電壓Vcs的最大值則由第二過電流保護閾值電壓Vth_min決定。 Fig. 13 is a view showing the relationship between the output frequency Fcv2 of the voltage controlled oscillator VCO and the voltage signal Vea2. As shown in FIG. 13, when the voltage signal Vea2 is smaller than the voltage Vg, the frequency Fcv2 of the level signal CV2 outputted by the voltage controlled oscillator VCO is the frequency Fmin, and the voltage control signal is oscillated when the voltage signal Vea2 is greater than the voltage Vf and smaller than the voltage Ve. The frequency Fcv2 of the level signal CV2 output by the VCO is Fcc_low, and when the voltage signal Vea2 changes between the voltage Vg and the voltage Vf, the frequency Fcv2 of the level signal CV2 outputted by the voltage controlled oscillator VCO gradually rises from the frequency Fmin to The frequency Fcc_low, when the voltage of the voltage signal Vea2 is greater than the voltage Ve, the level signal CV2 is always at a high level, that is, the frequency Fcv2 of the level signal CV2 is zero. Therefore, the frequency Fcv2 of the level signal CV2 also increases with the charging voltage Vo of the battery, from the frequency of 0 to the frequency Fcc_low, and then to the frequency Fmin. When the charging voltage Vo of the battery is equal to the voltage V2, the voltage signal Vea2 controls the voltage controlled oscillator VCO output frequency Fcv2 to be smaller than the frequency Fcc_low and the frequency Fcc_high, so the frequency Fcc_low and the frequency Fcc_high are masked, and the control signal s1 is determined by the level signal CV2, The maximum value of the voltage Vcs on the current detecting resistor Rs is determined by the second overcurrent protection threshold voltage Vth_min.

第14圖示出了根據本發明示例性實施例的根據電池的充電電壓Vo的變化而改變的電池的充電電流Io、過電流保護的電壓Vcs_peak以及開關Q的工作頻率(即反激式電源變換系統的工作頻率)Fo的變化的示意圖。 14 is a diagram showing a charging current Io of a battery, a voltage Vcs_peak of an overcurrent protection, and an operating frequency of a switch Q (ie, a flyback power conversion) that are changed according to a change in a charging voltage Vo of a battery according to an exemplary embodiment of the present invention. Schematic diagram of the change in the operating frequency of the system.

根據第14圖所示，當電池的充電電壓Vo小於電壓V1時，根據本發明示例性實施例的反激式電源變換系統工作在第一恒流充電控制區域，電池的充電電流為恒定的電流Icc_h，用於過電流保護的電壓Vcs_peak 為Vth_max，而開關Q的工作頻率由電平信號cc_high的頻率Fcc_high決定，並且該頻率隨著電壓的升高而逐漸上升。當電池的充電電壓Vo等於電壓V1時，所述反激式電源變換系統工作在第一恒壓充電控制區域，電池的充電電流Io逐漸下降至電流Icc_l，用於過電流保護的電壓Vcs_peak由第一過電流保護閾值電壓Vth_max逐漸降低變為第二過電流保護閾值電壓Vth_min，而開關Q的工作頻率如頻率Fcv1所示逐漸下降。當電池的充電電壓Vo大於電壓V1時，所述反激式電源變換系統工作在第二恒流充電控制區域，電池的充電電流Io為恒定的電流Icc_l(Icc_h大於Icc_l)，而用於過電流保護的電壓Vcs_peak為第二過電流保護閾值電壓Vth_min，而開關Q的工作頻率由電平信號CC_low的頻率Fcc_low決定，並且該頻率隨著電壓的升高而逐漸上升。當電池的充電電壓Vo等於電壓V2時，系統工作在第二恒壓充電控制區域，電池的充電電流Io逐漸下降至0，而開關Q的工作頻率如頻率Fcv2所示由頻率Fcc_low逐漸下降至最小頻率Fmin。 According to FIG. 14, when the charging voltage Vo of the battery is less than the voltage V1, the flyback power conversion system according to an exemplary embodiment of the present invention operates in the first constant current charging control region, and the charging current of the battery is a constant current. Icc_h, voltage for overcurrent protection Vcs_peak It is Vth_max, and the operating frequency of the switch Q is determined by the frequency Fcc_high of the level signal cc_high, and the frequency gradually rises as the voltage increases. When the charging voltage Vo of the battery is equal to the voltage V1, the flyback power conversion system operates in the first constant voltage charging control region, the charging current Io of the battery gradually decreases to the current Icc_l, and the voltage for the overcurrent protection Vcs_peak is An overcurrent protection threshold voltage Vth_max gradually decreases to become the second overcurrent protection threshold voltage Vth_min, and the operating frequency of the switch Q gradually decreases as indicated by the frequency Fcv1. When the charging voltage Vo of the battery is greater than the voltage V1, the flyback power conversion system operates in the second constant current charging control region, and the charging current Io of the battery is a constant current Icc_l (Icc_h is greater than Icc_l), and is used for overcurrent The protected voltage Vcs_peak is the second overcurrent protection threshold voltage Vth_min, and the operating frequency of the switch Q is determined by the frequency Fcc_low of the level signal CC_low, and the frequency gradually rises as the voltage increases. When the charging voltage Vo of the battery is equal to the voltage V2, the system operates in the second constant voltage charging control region, the charging current Io of the battery gradually drops to 0, and the operating frequency of the switch Q gradually decreases from the frequency Fcc_low to the minimum as indicated by the frequency Fcv2. Frequency Fmin.

為了方便本領域技術人員更好地理解開關Q的工作頻率Fo和電池的充電電壓Vo之間的關係，下面將對此進行簡略描述。 In order to facilitate the person skilled in the art to better understand the relationship between the operating frequency Fo of the switch Q and the charging voltage Vo of the battery, this will be briefly described below.

這裡，變壓器中電感和電壓、電流以及時間之間的關係為L．I=V．T，其中，L表示變壓器某一邊的電感量，I表示流過變壓器該邊的電流，V表示該邊的電壓，T表示退磁時間，因此，可以得到如下所示的公式： Here, the relationship between inductance and voltage, current and time in the transformer is L. I = V . T , where L represents the inductance of one side of the transformer, I represents the current flowing through the side of the transformer, V represents the voltage of the side, and T represents the demagnetization time, so that the formula shown below can be obtained:

在上面所述的公式裡，Tdemag表示退磁時間，Lsec表示副邊繞組的電感量，Isec表示副邊繞組的輸出電流，Vo表示電池的充電電壓，Lpri表示原邊的電感量，N表示原邊繞組和副邊繞組的匝數比，Ipri表示原邊繞組的電流值，Vcs_peak表示用於過電流保護的電壓，也即電阻Rs上施加的最高電壓。 In the formula described above, Tdemag represents the demagnetization time, Lsec represents the inductance of the secondary winding, Isec represents the output current of the secondary winding, Vo represents the charging voltage of the battery, Lpri represents the inductance of the primary side, and N represents the primary side. The turns ratio of the winding and the secondary winding, Ipri represents the current value of the primary winding, and Vcs_peak represents the voltage used for overcurrent protection, that is, the highest voltage applied across the resistor Rs.

以第一恒流充電控制模式為例，開關導通和關斷的週期Tcc_h=(1+k1)．Tdemag，綜合上述兩個公式，可知開關Q的工作頻率Fcc_h為： Taking the first constant current charging control mode as an example, the period of the switch on and off is Tcc _ h = (1 + k 1). Tdemag , combining the above two formulas, we know that the operating frequency Fcc_h of the switch Q is:

在第一恒流充電控制模式下，在電流檢測電阻Rs上輸出的電流(也即Vcs_peak/Rs)為定值(因為此時用於過電流保護的電壓Vcs_peak為第一過電流保護閾值電壓Vth_max)，Lpri，k1，N為定值，因此從上述公式可以知道，第一恒流充電控制模式下工作頻率Fcc_h和Vo成正比關係。同理，第二恒流充電控制模式下工作頻率Fcc_l和Vo也成正比關係。因此，在第一恒流充電控制模式和第二恒流充電控制模式下電池的充電電壓Vo越大，頻率越高。 In the first constant current charging control mode, the current outputted on the current detecting resistor Rs (ie, Vcs_peak/Rs) is a constant value (because the voltage Vcs_peak for overcurrent protection at this time is the first overcurrent protection threshold voltage Vth_max) ), Lpri, k1, N are constant values, so it can be known from the above formula that the operating frequency Fcc_h and Vo are proportional to each other in the first constant current charging control mode. Similarly, the operating frequency Fcc_l and Vo in the second constant current charging control mode are also proportional. Therefore, the larger the charging voltage Vo of the battery in the first constant current charging control mode and the second constant current charging control mode, the higher the frequency.

在第一恒壓充電控制模式和第二恒壓充電控制模式下，所述反激式電源變換系統的輸出功率Pout為： 這裡，Vcs_peak表示用於過電流保護的電壓。 In the first constant voltage charging control mode and the second constant voltage charging control mode, the output power Pout of the flyback power conversion system is: Here, Vcs_peak represents a voltage for overcurrent protection.

在第一恒壓充電控制模式下，電池的充電電壓Vo恒定，採樣電壓Vc0等於參考電壓Vref1，誤差放大器EA1輸出的電壓信號Vea1控制用 於過電流保護的電壓Vcs_peak，同時電壓信號Vea1也控制電平信號CV1的頻率。如果電壓信號Vea1的電壓越高，用於過電流保護的電壓Vcs_peak也越高，電平信號CV1的頻率(即開關Q的工作頻率Fo)越高，從上述公式也可以看出，輸出功率越大；反之，當電壓信號Vea1的電壓越低，輸出功率也越低。這裡，當採樣電壓Vc0高於或者低於參考電壓Vref1時，則誤差放大器EA1開環，電壓信號Vea1為低電平或者高電平，此時則不工作在第一恒壓充電控制模式。 In the first constant voltage charging control mode, the charging voltage Vo of the battery is constant, the sampling voltage Vc0 is equal to the reference voltage Vref1, and the voltage signal Vea1 output by the error amplifier EA1 is controlled. The overcurrent protection voltage Vcs_peak, while the voltage signal Vea1 also controls the frequency of the level signal CV1. If the voltage of the voltage signal Vea1 is higher, the voltage Vcs_peak for overcurrent protection is also higher, and the frequency of the level signal CV1 (ie, the operating frequency Fo of the switch Q) is higher. As can be seen from the above formula, the output power is higher. Large; conversely, the lower the voltage of the voltage signal Vea1, the lower the output power. Here, when the sampling voltage Vc0 is higher or lower than the reference voltage Vref1, the error amplifier EA1 is opened, and the voltage signal Vea1 is at a low level or a high level, and at this time, the first constant voltage charging control mode is not operated.

在第二恒壓充電控制模式下，電池的充電電壓Vo恒定，採樣電壓Vc0等於參考電壓Vref2，誤差放大器EA2輸出電壓信號Vea2來控制壓控振盪器VCO頻率輸出。此時，用於過電流保護的電壓Vcs_peak固定在第二過電流保護閾值電壓Vth_min，即電壓信號Vea2只控制公式中Fo頻率，電壓信號Vea2越高，壓控振盪器VCO輸出頻率(也即電平信號CV2的頻率Fcv2)越高，輸出功率越高，反之，輸出功率越低。這裡，Io和Fo成正比關係。 In the second constant voltage charging control mode, the charging voltage Vo of the battery is constant, the sampling voltage Vc0 is equal to the reference voltage Vref2, and the error amplifier EA2 outputs the voltage signal Vea2 to control the voltage controlled oscillator VCO frequency output. At this time, the voltage Vcs_peak for overcurrent protection is fixed at the second overcurrent protection threshold voltage Vth_min, that is, the voltage signal Vea2 only controls the Fo frequency in the formula, and the higher the voltage signal Vea2, the voltage controlled oscillator VCO output frequency (ie, electricity) The higher the frequency Fcv2) of the flat signal CV2, the higher the output power, and conversely, the lower the output power. Here, Io and Fo are proportional.

儘管已描述了本發明的特定實例，然而本領域技術人員應該明白，存在與所描述實例等同的其它實例。因此，本領域技術人員應該明白，本發明不局限於所示出的特定實例，而是僅由申請專利範圍的範圍來限定。 Although specific examples of the invention have been described, those skilled in the art will appreciate that there are other examples that are equivalent to the examples described. Therefore, it should be understood by those skilled in the art that the invention is not limited to the specific examples shown, but only by the scope of the claims.

510‧‧‧整流電路 510‧‧‧Rectifier circuit

520‧‧‧充電電路 520‧‧‧Charging circuit

530‧‧‧充電控制電路 530‧‧‧Charging control circuit

5310‧‧‧模式選擇電路 5310‧‧‧ mode selection circuit

5320‧‧‧邏輯驅動電路 5320‧‧‧Logical drive circuit

5330‧‧‧過電流保護電路 5330‧‧‧Overcurrent protection circuit

C1,C2‧‧‧電容 C1, C2‧‧‧ capacitor

D1,D3‧‧‧二極體 D1, D3‧‧‧ diode

DRV‧‧‧端子 DRV‧‧‧ terminal

Np‧‧‧原邊繞組 Np‧‧‧ primary winding

Nsec‧‧‧副邊繞組 Nsec‧‧‧ secondary winding

Naux‧‧‧輔助繞組 Naux‧‧‧Auxiliary winding

Q‧‧‧開關 Q‧‧‧ switch

R1,R2‧‧‧電阻 R1, R2‧‧‧ resistance

Rs‧‧‧電流檢測電阻 Rs‧‧‧ current sense resistor

Req‧‧‧輸出線等效電阻 Req‧‧‧output line equivalent resistance

s1,s2‧‧‧控制信號 S1, s2‧‧‧ control signal

Sdrv‧‧‧驅動控制信號 Sdrv‧‧‧ drive control signal

Vcs‧‧‧電壓/電壓信號 Vcs‧‧‧voltage/voltage signal

Vaux‧‧‧電壓 Vaux‧‧‧ voltage

Vctrl‧‧‧電壓控制信號 Vctrl‧‧‧ voltage control signal

Vin‧‧‧電壓信號 Vin‧‧‧ voltage signal

Vo‧‧‧充電電壓/輸出電壓 Vo‧‧‧Charging voltage/output voltage

Vth_max‧‧‧第一過電流保護閾值電壓 Vth_max‧‧‧First overcurrent protection threshold voltage

Vth_min‧‧‧第二過電流保護閾值電壓 Vth_min‧‧‧Second overcurrent protection threshold voltage